Audio encoder and decoder using a frequency domain processor with full-band gap filling and a time domain processor

ABSTRACT

An audio encoder for encoding an audio signal has: a first encoding processor for encoding a first audio signal portion in a frequency domain, having: a time frequency converter for converting the first audio signal portion into a frequency domain representation; an analyzer for analyzing the frequency domain representation to determine first spectral portions to be encoded with a first spectral resolution and second regions to be encoded with a second resolution; and a spectral encoder for encoding the first spectral portions with the first spectral resolution and encoding the second portions with the second resolution; a second encoding processor for encoding a second different audio signal portion in the time domain; a controller for analyzing and determining, which portion of the audio signal is the first audio signal portion encoded in the frequency domain and which portion is the second audio signal portion encoded in the time domain; and an encoded signal former for forming an encoded audio signal having a first encoded signal portion for the first audio signal portion and a second encoded signal portion for the second portion.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of co-pending U.S. application Ser.No. 18/158,035 filed Jan. 23, 2023 which is a continuation of co-pendingU.S. application Ser. No. 17/336,132 filed Jun. 1, 2021 which is acontinuation of co-pending U.S. application Ser. No. 16/286,397 filedFeb. 26, 2019 which is a continuation of co-pending U.S. applicationSer. No. 15/414,427 filed Jan. 24, 2017 which is now U.S. Pat. No.10,332,535 issued Jun. 25 2019; which is a continuation of InternationalApplication No. PCT/EP2015/067003, filed Jul. 24, 2015 and additionallyclaims priority from European Application No. 14178817.4, filed Jul. 28,2014, of which all the foregoing applications and patents areincorporated herein by reference in their entirety.

BACKGROUND OF THE INVENTION

The present invention relates to audio signal encoding and decoding and,in particular, to audio signal processing using parallel frequencydomain and time domain encoder/decoder processors.

The perceptual coding of audio signals for the purpose of data reductionfor efficient storage or transmission of these signals is a widely usedpractice. In particular when lowest bit rates are to be achieved, theemployed coding leads to a reduction of audio quality that often isprimarily caused by a limitation at the encoder side of the audio signalbandwidth to be transmitted. Here, typically the audio signal islow-pass filtered such that no spectral waveform content remains above acertain pre-determined cut-off frequency.

In contemporary codecs well-known methods exist for the decoder-sidesignal restoration through audio signal Bandwidth Extension (BWE), e.g.Spectral Band Replication (SBR) that operates in frequency domain orso-called Time Domain Bandwidth Extension (TD-BWE) being is apost-processor in speech coders that operates in time domain.

Additionally, several combined time domain/frequency domain codingconcepts exist such as concepts known under the term AMR-WB+ or USAC.

All these combined time domain/coding concepts have in common that thefrequency domain coder relies on bandwidth extension technologies whichincur a band limitation into the input audio signal and the portionabove a cross-over frequency or border frequency is encoded with a lowresolution coding concept and synthesized on the decoder-side. Hence,such concepts mainly rely on a pre-processor technology on the encoderside and a corresponding post-processing functionality on thedecoder-side.

Typically, the time domain encoder is selected for useful signals to beencoded in the time domain such as speech signals and the frequencydomain encoder is selected for non-speech signals, music signals, etc.However, specifically for non-speech signals having prominent harmonicsin the high frequency band, the known frequency domain encoders have areduced accuracy and, therefore, a reduced audio quality due to the factthat such prominent harmonics can only be separately parametricallyencoded or are eliminated at all in the encoding/decoding process.

Furthermore, concepts exist in which the time domain encoding/decodingbranch additionally relies on the bandwidth extension which alsoparametrically encodes an upper frequency range while a lower frequencyrange is typically encoded using an ACELP or any other CELP relatedcoder, for example a speech coder. This bandwidth extensionfunctionality increases the bitrate efficiency but, on the other hand,introduces further inflexibility due to the fact that both encodingbranches, i.e., the frequency domain encoding branch and the time domainencoding branch are band limited due to the bandwidth extensionprocedure or spectral band replication procedure operating above acertain crossover frequency substantially lower than the maximumfrequency included in the input audio signal

Relevant Topics in the State-of-Art Comprise

-   -   SBR as a post-processor to waveform decoding [1-3]    -   MPEG-D USAC core switching [4]    -   MPEG-H 3D IGF [5]

The following papers and patents describe methods that are considered toconstitute known technology for the application:

-   [1] M. Dietz, L. Liljeryd, K. Kjörling and O. Kunz, “Spectral Band    Replication, a novel approach in audio coding,” in 112th AES    Convention, Munich, Germany, 2002.-   [2] S. Meltzer, R. Bohm and F. Henn, “SBR enhanced audio codecs for    digital broadcasting such as “Digital Radio Mondiale” (DRM),” in    112th AES Convention, Munich, Germany, 2002.-   [3] T. Ziegler, A. Ehret, P. Ekstrand and M. Lutzky, “Enhancing mp3    with SBR: Features and Capabilities of the new mp3PRO Algorithm,” in    112th AES Convention, Munich, Germany, 2002.-   [4] MPEG-D USAC Standard.-   [5] PCT/EP2014/065109.

In MPEG-D USAC, a switchable core coder is described. However, in USAC,the band-limited core is restricted to transmit a low-pass filteredsignal. Therefore, certain music signals that contain prominent highfrequency content e.g. full-band sweeps, triangle sounds, etc. cannot bereproduced faithfully.

SUMMARY

According to an embodiment, an audio encoder for encoding an audiosignal may have: a first encoding processor for encoding a first audiosignal portion in a frequency domain, wherein the first encodingprocessor has: a time frequency converter for converting the first audiosignal portion into a frequency domain representation having spectrallines up to a maximum frequency of the first audio signal portion; ananalyzer for analyzing the frequency domain representation up to themaximum frequency to determine first spectral portions to be encodedwith a first spectral resolution and second spectral portions to beencoded with a second spectral resolution, the second spectralresolution being lower than the first spectral resolution, wherein theanalyzer is configured to determine a first spectral portion from thefirst spectral portions, the first spectral portion being placed, withrespect to frequency, between two second spectral portions from thesecond spectral portions; a spectral encoder for encoding the firstspectral portions with the first spectral resolution and for encodingthe second spectral portions with the second spectral resolution,wherein the spectral encoder has a parametric coder for calculatingspectral envelope information having the second spectral resolution fromthe second spectral portions; a second encoding processor for encoding asecond different audio signal portion in the time domain, wherein thesecond encoding processor has: a sampling rate converter for convertingthe second audio signal portion to a lower sampling rate representation,the lower sampling rate being lower than a sampling rate of the audiosignal, wherein the lower sampling rate representation does not includethe high band of the input signal; a time domain low band encoder fortime domain encoding the lower sampling rate representation; and a timedomain bandwidth extension encoder for parametrically encoding the highband; a controller configured for analyzing the audio signal and fordetermining, which portion of the audio signal is the first audio signalportion encoded in the frequency domain and which portion of the audiosignal is the second audio signal portion encoded in the time domain;and an encoded signal former for forming an encoded audio signal havinga first encoded signal portion for the first audio signal portion and asecond encoded signal portion for the second audio signal portion.

According to another embodiment, an audio decoder for decoding anencoded audio signal may have: a first decoding processor for decoding afirst encoded audio signal portion in a frequency domain, the firstdecoding processor having: a spectral decoder for decoding firstspectral portions with a high spectral resolution and for synthesizingsecond spectral portions using a parametric representation of the secondspectral portions and at least a decoded first spectral portion toobtain a decoded spectral representation, wherein the spectral decoderis configured to generate the first decoded representation so that afirst spectral portion is placed with respect to frequency between twosecond spectral portions; and a frequency-time converter for convertingthe decoded spectral representation into a time domain to obtain adecoded first audio signal portion; a second decoding processor fordecoding a second encoded audio signal portion in the time domain toobtain a decoded second audio signal portion, wherein the seconddecoding processor has: a time domain low band decoder for decoding alow band time domain signal; an upsampler for upsampling the low bandtime domain signal; a time domain bandwidth extension decoder forsynthesizing a high band of a time domain output signal; and a mixer formixing a synthesized high band of the time domain signal and anupsampled low band time domain signal; and a combiner for combining thedecoded first spectral portion and the decoded second spectral portionto obtain a decoded audio signal.

According to still another embodiment, a method of encoding an audiosignal may have the steps of: first encoding a first audio signalportion in a frequency domain, wherein the first encoding has:converting the first audio signal portion into a frequency domainrepresentation having spectral lines up to a maximum frequency of thefirst audio signal portion; analyzing the frequency domainrepresentation up to the maximum frequency to determine first spectralportions to be encoded with a first spectral resolution and secondspectral portions to be encoded with a second spectral resolution, thesecond spectral resolution being lower than the first spectralresolution, wherein the analyzing determines a first spectral portionfrom the first spectral portions, the first spectral portion beingplaced, with respect to frequency, between two second spectral portionsfrom the second spectral portions; encoding the first spectral portionswith the first spectral resolution and for encoding the second spectralportions with the second spectral resolution, wherein the encoding thesecond spectral portion has calculating, from the second spectralportions, spectral envelope information having the second spectralresolution; second encoding a second different audio signal portion inthe time domain wherein the second encoding has: converting the secondaudio signal portion to a lower sampling rate representation, the lowersampling rate being lower than a sampling rate of the audio signal,wherein the lower sampling rate representation does not include the highband of the input signal; time domain encoding the lower sampling raterepresentation; and parametrically encoding the high band; analyzing theaudio signal and determining, which portion of the audio signal is thefirst audio signal portion encoded in the frequency domain and whichportion of the audio signal is the second audio signal portion encodedin the time domain; and forming an encoded audio signal having a firstencoded signal portion for the first audio signal portion and a secondencoded signal portion for the second audio signal portion.

According to another embodiment, a method of decoding an encoded audiosignal may have the steps of: first decoding a first encoded audiosignal portion in a frequency domain, the first decoding having:decoding first spectral portions with a high spectral resolution andsynthesizing second spectral portions using a parametric representationof the second spectral portions and at least a decoded first spectralportion to obtain a decoded spectral representation, wherein decodinghas generating the first decoded representation so that a first spectralportion is placed with respect to frequency between two second spectralportions; and converting the decoded spectral representation into a timedomain to obtain a decoded first audio signal portion; second decoding asecond encoded audio signal portion in the time domain to obtain adecoded second audio signal portion, wherein the second decoding has:decoding a low band time domain signal; upsampling the low band timedomain signal; synthesizing a high band of a time domain output signal;and mixing a synthesized high band of the time domain signal and anupsampled low band time domain signal; and combining the decoded firstspectral portion and the decoded second spectral portion to obtain adecoded audio signal.

Another embodiment may have a non-transitory digital storage mediumhaving stored thereon a computer program for performing a method ofencoding an audio signal, having: first encoding a first audio signalportion in a frequency domain, wherein the first encoding has:converting the first audio signal portion into a frequency domainrepresentation having spectral lines up to a maximum frequency of thefirst audio signal portion; analyzing the frequency domainrepresentation up to the maximum frequency to determine first spectralportions to be encoded with a first spectral resolution and secondspectral portions to be encoded with a second spectral resolution, thesecond spectral resolution being lower than the first spectralresolution, wherein the analyzing determines a first spectral portionfrom the first spectral portions, the first spectral portion beingplaced, with respect to frequency, between two second spectral portionsfrom the second spectral portions; encoding the first spectral portionswith the first spectral resolution and for encoding the second spectralportions with the second spectral resolution, wherein the encoding thesecond spectral portion has calculating, from the second spectralportions, spectral envelope information having the second spectralresolution; second encoding a second different audio signal portion inthe time domain wherein the second encoding has: converting the secondaudio signal portion to a lower sampling rate representation, the lowersampling rate being lower than a sampling rate of the audio signal,wherein the lower sampling rate representation does not include the highband of the input signal; time domain encoding the lower sampling raterepresentation; and parametrically encoding the high band; analyzing theaudio signal and determining, which portion of the audio signal is thefirst audio signal portion encoded in the frequency domain and whichportion of the audio signal is the second audio signal portion encodedin the time domain; and forming an encoded audio signal having a firstencoded signal portion for the first audio signal portion and a secondencoded signal portion for the second audio signal portion, when saidcomputer program is run by a computer.

Still another embodiment may have a non-transitory digital storagemedium having stored thereon a computer program for performing a methodof decoding an encoded audio signal, having: first decoding a firstencoded audio signal portion in a frequency domain, the first decodinghaving: decoding first spectral portions with a high spectral resolutionand synthesizing second spectral portions using a parametricrepresentation of the second spectral portions and at least a decodedfirst spectral portion to obtain a decoded spectral representation,wherein decoding has generating the first decoded representation so thata first spectral portion is placed with respect to frequency between twosecond spectral portions; and converting the decoded spectralrepresentation into a time domain to obtain a decoded first audio signalportion; second decoding a second encoded audio signal portion in thetime domain to obtain a decoded second audio signal portion, wherein thesecond decoding has: decoding a low band time domain signal; upsamplingthe low band time domain signal; synthesizing a high band of a timedomain output signal; and mixing a synthesized high band of the timedomain signal and an upsampled low band time domain signal; andcombining the decoded first spectral portion and the decoded secondspectral portion to obtain a decoded audio signal, when said computerprogram is run by a computer.

The present invention is based on the finding that a time domainencoding/decoding processor can be combined with a frequency domainencoding/decoding processor having a gap filling functionality but thisgap filling functionality for filling spectral holes is operated overthe whole band of the audio signal or at least above a certain gapfilling frequency. Importantly, the frequency domain encoding/decodingprocessor is particularly in the position to perform accurate or waveform or spectral value encoding/decoding up to the maximum frequency andnot only until a crossover frequency. Furthermore, the full-bandcapability of the frequency domain encoder for encoding with the highresolution allows an integration of the gap filling functionality intothe frequency domain encoder.

Hence, in accordance with the present invention by using the full-bandspectral encoder/decoder processor, the problems related to theseparation of the bandwidth extension on the one hand and the corecoding on the other hand can be addressed and overcome by performing thebandwidth extension in the same spectral domain in which the coredecoder operates. Therefore, a full rate core decoder is provided whichencodes and decodes the full audio signal range. This does not requirethe need for a downsampler on the encoder side and an upsampler on thedecoder side. Instead, the whole processing is performed in the fullsampling rate or full-bandwidth domain. In order to obtain a high codinggain, the audio signal is analyzed in order to find a first set of firstspectral portions which has to be encoded with a high resolution, wherethis first set of first spectral portions may include, in an embodiment,tonal portions of the audio signal. On the other hand, non-tonal ornoisy components in the audio signal constituting a second set of secondspectral portions are parametrically encoded with low spectralresolution. The encoded audio signal then only necessitates the firstset of first spectral portions encoded in a waveform-preserving mannerwith a high spectral resolution and, additionally, the second set ofsecond spectral portions encoded parametrically with a low resolutionusing frequency “tiles” sourced from the first set. On the decoder side,the core decoder, which is a full-band decoder, reconstructs the firstset of first spectral portions in a waveform-preserving manner, i.e.,without any knowledge that there is any additional frequencyregeneration. However, the so generated spectrum has a lot of spectralgaps. These gaps are subsequently filled with the inventive IntelligentGap Filling (IGF) technology by using a frequency regeneration applyingparametric data on the one hand and using a source spectral range, i.e.,first spectral portions reconstructed by the full rate audio decoder onthe other hand.

In further embodiments, spectral portions, which are reconstructed bynoise filling only rather than bandwidth replication or frequency tilefilling, constitute a third set of third spectral portions. Due to thefact that the coding concept operates in a single domain for the corecoding/decoding on the one hand and the frequency regeneration on theother hand, the IGF is not only restricted to fill up a higher frequencyrange but can fill up lower frequency ranges, either by noise fillingwithout frequency regeneration or by frequency regeneration using afrequency tile at a different frequency range.

Furthermore, it is emphasized that an information on spectral energies,an information on individual energies or an individual energyinformation, an information on a survive energy or a survive energyinformation, an information a tile energy or a tile energy information,or an information on a missing energy or a missing energy informationmay comprise not only an energy value, but also an (e.g. absolute)amplitude value, a level value or any other value, from which a finalenergy value can be derived. Hence, the information on an energy maye.g. comprise the energy value itself, and/or a value of a level and/orof an amplitude and/or of an absolute amplitude.

A further aspect is based on the finding that the correlation situationis not only important for the source range but is also important for thetarget range. Furthermore, the present invention acknowledges thesituation that different correlation situations can occur in the sourcerange and the target range. When, for example, a speech signal with highfrequency noise is considered, the situation can be that the lowfrequency band comprising the speech signal with a small number ofovertones is highly correlated in the left channel and the rightchannel, when the speaker is placed in the middle. The high frequencyportion, however, can be strongly uncorrelated due to the fact thatthere might be a different high frequency noise on the left sidecompared to another high frequency noise or no high frequency noise onthe right side. Thus, when a straightforward gap filling operation wouldbe performed that ignores this situation, then the high frequencyportion would be correlated as well, and this might generate seriousspatial segregation artifacts in the reconstructed signal. In order toaddress this issue, parametric data for a reconstruction band or,generally, for the second set of second spectral portions which have tobe reconstructed using a first set of first spectral portions iscalculated to identify either a first or a second different two-channelrepresentation for the second spectral portion or, stated differently,for the reconstruction band. On the encoder side, a two-channelidentification is, therefore calculated for the second spectralportions, i.e., for the portions, for which, additionally, energyinformation for reconstruction bands is calculated. A frequencyregenerator on the decoder side then regenerates a second spectralportion depending on a first portion of the first set of first spectralportions, i.e., the source range and parametric data for the secondportion such as spectral envelope energy information or any otherspectral envelope data and, additionally, dependent on the two-channelidentification for the second portion, i.e., for this reconstructionband under reconsideration.

The two-channel identification is advantageously transmitted as a flagfor each reconstruction band and this data is transmitted from anencoder to a decoder and the decoder then decodes the core signal asindicated by advantageously calculated flags for the core bands. Then,in an implementation, the core signal is stored in both stereorepresentations (e.g. left/right and mid/side) and, for the IGFfrequency tile filling, the source tile representation is chosen to fitthe target tile representation as indicated by the two-channelidentification flags for the intelligent gap filling or reconstructionbands, i.e., for the target range.

It is emphasized that this procedure not only works for stereo signals,i.e., for a left channel and the right channel but also operates formulti-channel signals. In the case of multi-channel signals, severalpairs of different channels can be processed in that way such as a leftand a right channel as a first pair, a left surround channel and a rightsurround as the second pair and a center channel and an LFE channel asthe third pair. Other pairings can be determined for higher outputchannel formats such as 7.1, 11.1 and so on.

A further aspect is based on the finding that the audio quality of thereconstructed signal can be improved through IGF since the wholespectrum is accessible to the core encoder so that, for example,perceptually important tonal portions in a high spectral range can stillbe encoded by the core coder rather than parametric substitution.Additionally, a gap filling operation using frequency tiles from a firstset of first spectral portions which is, for example, a set of tonalportions typically from a lower frequency range, but also from a higherfrequency range if available, is performed. For the spectral envelopeadjustment on the decoder side, however, the spectral portions from thefirst set of spectral portions located in the reconstruction band arenot further post-processed by e.g. the spectral envelope adjustment.Only the remaining spectral values in the reconstruction band which donot originate from the core decoder are to be envelope adjusted usingenvelope information. Advantageously, the envelope information is afull-band envelope information accounting for the energy of the firstset of first spectral portions in the reconstruction band and the secondset of second spectral portions in the same reconstruction band, wherethe latter spectral values in the second set of second spectral portionsare indicated to be zero and are, therefore, not encoded by the coreencoder, but are parametrically coded with low resolution energyinformation.

It has been found that absolute energy values, either normalized withrespect to the bandwidth of the corresponding band or not normalized,are useful and very efficient in an application on the decoder side.This especially applies when gain factors have to be calculated based ona residual energy in the reconstruction band, the missing energy in thereconstruction band and frequency tile information in the reconstructionband.

Furthermore, it is of advantage that the encoded bitstream not onlycovers energy information for the reconstruction bands but,additionally, scale factors for scale factor bands extending up to themaximum frequency. This ensures that for each reconstruction band, forwhich a certain tonal portion, i.e., a first spectral portion isavailable, this first set of first spectral portion can actually bedecoded with the right amplitude. Furthermore, in addition to the scalefactor for each reconstruction band, an energy for this reconstructionband is generated in an encoder and transmitted to a decoder.Furthermore, it is of advantage that the reconstruction bands coincidewith the scale factor bands or in case of energy grouping, at least theborders of a reconstruction band coincide with borders of scale factorbands.

A further aspect is based on the finding that certain impairments inaudio quality can be remedied by applying a signal adaptive frequencytile filling scheme. To this end, an analysis on the encoder-side isperformed in order to find out the best matching source region candidatefor a certain target region. A matching information identifying for atarget region a certain source region together with optionally someadditional information is generated and transmitted as side informationto the decoder. The decoder then applies a frequency tile fillingoperation using the matching information. To this end, the decoder readsthe matching information from the transmitted data stream or data fileand accesses the source region identified for a certain reconstructionband and, if indicated in the matching information, additionallyperforms some processing of this source region data to generate rawspectral data for the reconstruction band. Then, this result of thefrequency tile filling operation, i.e., the raw spectral data for thereconstruction band, is shaped using spectral envelope information inorder to finally obtain a reconstruction band that comprises the firstspectral portions such as tonal portions as well. These tonal portions,however, are not generated by the adaptive tile filling scheme, butthese first spectral portions are output by the audio decoder or coredecoder directly.

The adaptive spectral tile selection scheme may operate with a lowgranularity. In this implementation, a source region is subdivided intotypically overlapping source regions and the target region or thereconstruction bands are given by non-overlapping frequency targetregions. Then, similarities between each source region and each targetregion are determined on the encoder-side and the best matching pair ofa source region and the target region are identified by the matchinginformation and, on the decoder-side, the source region identified inthe matching information is used for generating the raw spectral datafor the reconstruction band.

For the purpose of obtaining a higher granularity, each source region isallowed to shift in order to obtain a certain lag where the similaritiesare maximum. This lag can be as fine as a frequency bin and allows aneven better matching between a source region and the target region.

Furthermore, in addition of only identifying a best matching pair, thiscorrelation lag can also be transmitted within the matching informationand, additionally, even a sign can be transmitted. When the sign isdetermined to be negative on the encoder-side, then a corresponding signflag is also transmitted within the matching information and, on thedecoder-side, the source region spectral values are multiplied by “−1”or, in a complex representation, are “rotated” by 180 degrees.

A further implementation of this invention applies a tile whiteningoperation. Whitening of a spectrum removes the coarse spectral envelopeinformation and emphasizes the spectral fine structure which is offoremost interest for evaluating tile similarity. Therefore, a frequencytile on the one hand and/or the source signal on the other hand arewhitened before calculating a cross correlation measure. When only thetile is whitened using a predefined procedure, a whitening flag istransmitted indicating to the decoder that the same predefined whiteningprocess shall be applied to the frequency tile within IGF.

Regarding the tile selection, it is of advantage to use the lag of thecorrelation to spectrally shift the regenerated spectrum by an integernumber of transform bins. Depending on the underlying transform, thespectral shifting may necessitate addition corrections. In case of oddlags, the tile is additionally modulated through multiplication by analternating temporal sequence of −1/1 to compensate for thefrequency-reversed representation of every other band within the MDCT.Furthermore, the sign of the correlation result is applied whengenerating the frequency tile.

Furthermore, it is of advantage to use tile pruning and stabilization inorder to make sure that artifacts created by fast changing sourceregions for the same reconstruction region or target region are avoided.To this end, a similarity analysis among the different identified sourceregions is performed and when a source tile is similar to other sourcetiles with a similarity above a threshold, then this source tile can bedropped from the set of potential source tiles since it is highlycorrelated with other source tiles. Furthermore, as a kind of tileselection stabilization, it is of advantage to keep the tile order fromthe previous frame if none of the source tiles in the current framecorrelate (better than a given threshold) with the target tiles in thecurrent frame.

A further aspect is based on the finding that an improved quality andreduced bitrate specifically for signals comprising transient portionsas they occur very often in audio signals is obtained by combining theTemporal Noise Shaping (TNS) or Temporal Tile Shaping (TTS) technologywith high frequency reconstruction. The TNS/TTS processing on theencoder-side being implemented by a prediction over frequencyreconstructs the time envelope of the audio signal. Depending on theimplementation, i.e., when the temporal noise shaping filter isdetermined within a frequency range not only covering the sourcefrequency range but also the target frequency range to be reconstructedin a frequency regeneration decoder, the temporal envelope is not onlyapplied to the core audio signal up to a gap filling start frequency,but the temporal envelope is also applied to the spectral ranges ofreconstructed second spectral portions. Thus, pre-echoes or post-echoesthat would occur without temporal tile shaping are reduced oreliminated. This is accomplished by applying an inverse prediction overfrequency not only within the core frequency range up to a certain gapfilling start frequency but also within a frequency range above the corefrequency range. To this end, the frequency regeneration or frequencytile generation is performed on the decoder-side before applying aprediction over frequency. However, the prediction over frequency caneither be applied before or subsequent to spectral envelope shapingdepending on whether the energy information calculation has beenperformed on the spectral residual values subsequent to filtering or tothe (full) spectral values before envelope shaping.

The TTS processing over one or more frequency tiles additionallyestablishes a continuity of correlation between the source range and thereconstruction range or in two adjacent reconstruction ranges orfrequency tiles.

In an implementation, it is of advantage to use complex TNS/TTSfiltering. Thereby, the (temporal) aliasing artifacts of a criticallysampled real representation, like MDCT, are avoided. A complex TNSfilter can be calculated on the encoder-side by applying not only amodified discrete cosine transform but also a modified discrete sinetransform in addition to obtain a complex modified transform.Nevertheless, only the modified discrete cosine transform values, i.e.,the real part of the complex transform is transmitted. On thedecoder-side, however, it is possible to estimate the imaginary part ofthe transform using MDCT spectra of preceding or subsequent frames sothat, on the decoder-side, the complex filter can be again applied inthe inverse prediction over frequency and, specifically, the predictionover the border between the source range and the reconstruction rangeand also over the border between frequency-adjacent frequency tileswithin the reconstruction range.

The inventive audio coding system efficiently codes arbitrary audiosignals at a wide range of bitrates. Whereas, for high bitrates, theinventive system converges to transparency, for low bitrates perceptualannoyance is minimized. Therefore, the main share of available bitrateis used to waveform code just the perceptually most relevant structureof the signal in the encoder, and the resulting spectral gaps are filledin the decoder with signal content that roughly approximates theoriginal spectrum. A very limited bit budget is consumed to control theparameter driven so-called spectral Intelligent Gap Filling (IGF) bydedicated side information transmitted from the encoder to the decoder.

In further embodiments, the time domain encoding/decoding processorrelies on a lower sampling rate and the corresponding bandwidthextension functionality.

In further embodiments, a cross-processor is provided in order toinitialize the time domain encoder/decoder with initialization dataderived from the currently processed frequency domain encoder/decodersignal This allows that when the currently processed audio signalportion is processed by the frequency domain encoder, the parallel timedomain encoder is initialized so that when a switch from the frequencydomain encoder to a time domain encoder takes place, this time domainencoder can start processing since all the initialization data relatingto earlier signals are already there due to the cross-processor. Thiscross-processor may be applied on the encoder-side and, additionally, onthe decoder-side and may use a frequency-time transform whichadditionally performs a very efficient downsampling from the higheroutput or input sampling rate into the lower time domain core codersampling rate by only selecting a certain low band portion of the domainsignal together with a certain reduced transform size. Thus, a samplerate conversion from the high sampling rate to the low sampling rate isvery efficiently performed and this signal obtained by the transformwith the reduced transform size can then be used for initializing thetime domain encoder/decoder so that the time domain encoder/decoder isready to immediately perform time domain encoding when this situation issignaled by a controller and the immediately preceding audio signalportion was encoded in the frequency domain.

Hence, embodiments of the present invention allow a seamless switchingof a perceptual audio coder comprising spectral gap filling and a timedomain encoder with or without bandwidth extension.

Hence, the present invention relies on methods that are not restrictedto removing the high frequency content above a cut-off frequency in thefrequency domain encoder from the audio signal but rathersignal-adaptively removes spectral band-pass regions leaving spectralgaps in the encoder and subsequently reconstructs these spectral gaps inthe decoder. Advantageously, an integrated solution such as intelligentgap filling is used that efficiently combines full-bandwidth audiocoding and spectral gap filling particularly in the MDCT transformdomain.

Hence, the present invention provides an improved concept for combiningspeech coding and a subsequent time domain bandwidth extension with afull-band wave form decoding comprising spectral gap filling into aswitchable perceptual encoder/decoder.

Hence, in contrast to already existing methods, the new concept utilizesfull-band audio signal wave form coding in the transform domain coderand at the same time allows a seamless switching to a speech coderadvantageously followed by a time domain bandwidth extension.

Further embodiments of the present invention avoid the explainedproblems that occur due to a fixed band limitation. The concept enablesthe switchable combination of a full-band wave form coder in thefrequency domain equipped with a spectral gap filling and a lowersampling rate speech coder and a time domain bandwidth extension. Such acoder is capable of wave form coding the aforementioned problematicsignals providing full audio bandwidth up to the Nyquist frequency ofthe audio input signal. Nevertheless, seamless switching between bothcoding strategies is guaranteed particularly by the embodiments havingthe cross-processor. For this seamless switching, the cross-processorrepresents a cross connection at both encoder and decoder between thefull-band capable full-rate (input sampling rate) frequency domainencoder and the low-rate ACELP coder having a lower sampling rate toproperly initialize the ACELP parameters and buffers particularly withinthe adaptive codebook, the LPC filter or the resampling stage, whenswitching from the frequency domain coder such as TCX to the time domainencoder such as ACELP.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will subsequently be discussed withrespect to the accompanying drawings in which:

FIG. 1A illustrates an apparatus for encoding an audio signal;

FIG. 1B illustrates a decoder for decoding an encoded audio signalmatching with the encoder of FIG. 1A;

FIG. 2A illustrates an implementation of the decoder;

FIG. 2B illustrates an implementation of the encoder;

FIG. 3A illustrates a schematic representation of a spectrum asgenerated by the spectral domain decoder of FIG. 1B;

FIG. 3B illustrates a table indicating the relation between scalefactors for scale factor bands and energies for reconstruction bands andnoise filling information for a noise filling band;

FIG. 4A illustrates the functionality of the spectral domain encoder forapplying the selection of spectral portions into the first and secondsets of spectral portions;

FIG. 4B illustrates an implementation of the functionality of FIG. 4A;

FIG. 5A illustrates a functionality of an MDCT encoder;

FIG. 5B illustrates a functionality of the decoder with an MDCTtechnology;

FIG. 5C illustrates an implementation of the frequency regenerator;

FIG. 6 illustrates an implementation of an audio encoder;

FIG. 7A illustrates a cross-processor within the audio encoder;

FIG. 7B illustrates an implementation of an inverse or frequency-timetransform additionally providing a sampling rate reduction within thecross-processor;

FIG. 8 illustrates an implementation of the controller of FIG. 6 ;

FIG. 9 illustrates a further embodiment of the time domain encoderhaving bandwidth extension functionalities;

FIG. 10 illustrates an advantageous usage of a preprocessor;

FIG. 11A illustrates a schematic implementation of the audio decoder;

FIG. 11B illustrates a cross-processor within the decoder for providinginitialization data for the time domain decoder;

FIG. 12 illustrates an implementation of the time domain decodingprocessor of FIG. 11A;

FIG. 13 illustrates a further implementation of the time domainbandwidth extension;

FIG. 14A consisting of FIGS. 14A-1 and 14A-2 illustrates animplementation of an audio encoder;

FIG. 14B illustrates an implementation of an audio decoder; and

FIG. 14C illustrates an inventive implementation of a time domaindecoder with sample rate conversion and bandwidth extension.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 6 illustrates an audio encoder for encoding an audio signalcomprising a first encoding processor 600 for encoding a first audiosignal portion in a frequency domain. The first encoding processor 600comprises a time frequency converter 602 for converting the first inputaudio signal portion into a frequency domain representation havingspectral lines up to a maximum frequency of the input signal.Furthermore, the first encoding processor 600 comprises an analyzer 604for analyzing the frequency domain representation up to the maximumfrequency to determine first spectral regions to be encoded with a firstspectral representation and to determine second spectral regions to beencoded with a second spectral resolution being lower than the firstspectral resolution. In particular, the full-band analyzer 604determines which frequency lines or spectral values in the timefrequency converter spectrum are to be encoded spectral-line wise andwhich other spectral portions are to be encoded in a parametric way andthese latter spectral values are then reconstructed on the decoder-sidewith the gap filling procedure. The actual encoding operation isperformed by a spectral encoder 606 for encoding the first spectralregions or spectral portions with the first resolution and forparametrically encoding the second spectral regions or portions with thesecond spectral resolution.

The audio encoder of FIG. 6 additionally comprises a second encodingprocessor 610 for encoding the audio signal portion in a time domain.Additionally, the audio encoder comprises a controller 620 configuredfor analyzing the audio signal at an audio signal input 601 and fordetermining which portion of the audio signal is the first audio signalportion encoded in the frequency domain and which portion of the audiosignal is the second audio signal portion encoded in the time domain.Furthermore, an encoded signal former 630 which can be, for example,implemented as a bit stream multiplexor is provided which is configuredfor forming an encoded audio signal comprising a first encoded signalportion for the first audio signal portion and a second encoded signalportion for the second audio signal portion. Importantly, the encodedsignal only has either a frequency domain representation or a timedomain representation from one and the same audio signal portion.

Hence, the controller 620 makes sure that for a single audio signalportion only a time domain representation or a frequency domainrepresentation is in the encoded signal. This can be accomplished by thecontroller 620 in several ways. One way would be that, for one and thesame audio signal portion, both representations arrive at block 630 andthe controller 620 controls the encoded signal former 630 to onlyintroduce one of both representations into the encoded signal.Alternatively, however, the controller 620 can control an input into thefirst encoding processor and an input into the second encoding processorso that, based on the analysis of the corresponding signal portion, onlyone of both blocks 600 or 610 is activated to actually perform the fullencoding operation and the other block is deactivated.

This deactivation can be a deactivation or, as illustrated with respectto, for example, FIG. 7A, is only a kind of “initialization” mode wherethe other encoding processor is only active to receive and processinitialization data in order to initialize internal memories but anyspecific encoding operation is not performed at all. This activation canbe done by a certain switch at the input which is not illustrated inFIG. 6 or, advantageously, by control lines 621 and 622. Hence, in thisembodiment, the second encoding processor 610 does not output anythingwhen the controller 620 has determined that the current audio signalportion should be encoded by the first encoding processor but the secondencoding processor is nevertheless provided with initialization data tobe active for an instant switching in the future. On the other hand, thefirst encoding processor is configured to not need any data from thepast to update any internal memories and, therefore, when the currentaudio signal portion is to be encoded by the second encoding processor610 then the controller 620 can control the first ending encodingprocessor 600 via control line 621 to be inactive at all. This meansthat the first encoding processor 600 does not need to be in aninitialization state or waiting state but can be in a completedeactivation state. This is of advantage particularly for mobile deviceswhere power consumption and, therefore, battery life is an issue.

In the further specific implementation of the second encoding processoroperating in the time domain, the second encoding processor comprises adownsampler 900 or sampling rate converter for converting the audiosignal portion into a representation with a lower sampling rate, whereinthe lower sampling rate is lower than a sampling rate at the input intothe first encoding processor. This is illustrated in FIG. 9 . Inparticular, when the input audio signal comprises a low band and a highband, it is of advantage that the lower sampling rate representation atthe output of block 900 only has the low band of the input audio signalportion and this low band is then encoded by a time domain low bandencoder 910 which is configured for time-domain encoding the lowersampling rate representation provided by block 900. Furthermore, a timedomain bandwidth extension encoder 920 is provided for parametricallyencoding the high band. To this end, the time domain bandwidth extensionencoder 920 receives at least the high band of the input audio signal orthe low band and the high band of the input audio signal.

In a further embodiment of the present invention the audio encoderadditionally comprises, although not illustrated in FIG. 6 butillustrated in FIG. 10 , a preprocessor 1000 configured forpreprocessing the first audio signal portion and the second audio signalportion. In an embodiment, this preprocessor comprises a predictionanalyzer for determining prediction coefficients. This predictionanalyzer can be implemented as an LPC (linear prediction coding)analyzer for determining LPC coefficients. However, other analyzers canbe implemented as well. Furthermore, the preprocessor, which is alsoillustrated in FIG. 14A, comprises a prediction coefficient quantizer1010, wherein this device illustrated in FIG. 14A receives predictioncoefficient data from the prediction analyzer also illustrated in FIG.14A at 1002.

Furthermore, the preprocessor additionally comprises an entropy coderfor generating an encoded version of the quantized predictioncoefficients. It is important to note that the encoded signal former 630or the specific implementation, i.e., the bit stream multiplexor 613makes sure that the encoded version of the quantized predictioncoefficients is included into the encoded audio signal 632.Advantageously, the LPC coefficients are not directly quantized but areconverted into an ISF, for example, or any other representation bettersuited for quantization. This conversion may be performed either by thedetermined LPC coefficients block 1002 or is performed within the block1010 for quantizing the LPC coefficients.

Furthermore, the preprocessor may comprise a resampler 1004 forresampling an audio input signal at an input sampling rate into a lowersampling rate for the time domain encoder. When the time domain encoderis an ACELP encoder having a certain ACELP sampling rate then the downsampling is performed to advantageously either 12.8 kHz or 16 kHz. Theinput sampling rate can be any of a particular number of sampling ratessuch as 32 kHz or an even higher sampling rate. On the other hand, thesampling rate of the time domain encoder will be predetermined bycertain restrictions and the resampler 1004 performs this resampling andoutputs the lower sampling rate representation of the input signal.Hence, the resampler 1004 can perform a similar functionality and caneven be one and the same element as the downsampler 900 illustrated inthe context of FIG. 9 .

Furthermore, it is of advantage to apply a pre-emphasis in thepre-emphasis block 1005 in FIG. 14A. The pre-emphasis processing iswell-known in the art of time domain encoding and is described inliterature referring to the AMR-WB+ processing and the pre-emphasis isparticularly configured for compensating for a spectral tilt and,therefore, allows a better calculation of LPC parameters at a given LPCorder.

Furthermore, the preprocessor may additionally comprise a TCX-LTPparameter extraction for controlling an LTP post filter illustrated at1420 in FIG. 14B. This block is illustrated at 1006 in FIG. 14A.Furthermore, the preprocessor may additionally comprise otherfunctionalities illustrated at 1007 and these other functionalities maycomprise a pitch search functionality, a voice activity detection (VAD)functionality or any other functionalities known in the art of timedomain or speech coding.

As illustrated, the result of block 1006 is input into the encodedsignal, i.e., is in the embodiment of FIG. 14A, input into the bitstream multiplexor 630. Furthermore, if necessitated, data from block1007 can also be introduced into the bit stream multiplexor or can,alternatively, be used for the purpose of time domain encoding in thetime domain encoder.

Hence, to summarize, common to both paths is a preprocessing operation1000 in which commonly used signal processing operations are performed.These comprise a resampling to an ACELP sampling rate (12.8 or 16 kHz)for one parallel path and this resampling is performed. Furthermore, aTCX LTP parameter extraction illustrated at block 1006 is performed and,additionally, a pre-emphasis and a determination of LPC coefficients isperformed. As outlined, the pre-emphasis compensates for the spectraltilt and, therefore, makes the calculation of LPC parameters at a givenLPC order more efficient.

Subsequently, reference is made to FIG. 8 in order to illustrate animplementation of the controller 620. The controller receives, at aninput, the audio signal portion under consideration. Advantageously, asillustrated in 14A, the controller receives any signal available in thepreprocessor 1000 which can either be the original input signal at theinput sampling rate or a resampled version at the lower time domainencoder sampling rate or a signal obtained subsequent to thepre-emphasis processing in block 1005.

Based on this audio signal portion, the controller 620 addresses afrequency domain encoder simulator 621 and a time domain encodersimulator 622 in order to calculate for each encoder possibility anestimated signal to noise ratio. Subsequently, the selector 623 selectsthe encoder which has provided the better signal to noise ratio,naturally under the consideration of a predefined bit rate. The selectorthen identifies the corresponding encoder via the control output. Whenit is determined that the audio signal portion under consideration is tobe encoded using the frequency domain encoder, the time domain encoderis set into an initialization state or in other embodiments notrequiring a very instant switching in a completely deactivated state.However, when it is determined that the audio signal portion underconsideration is to be encoded by the time domain encoder, the frequencydomain encoder is then deactivated.

Subsequently, an implementation of the controller illustrated in FIG. 8is illustrated. The decision whether ACELP or TCX path should be chosenis performed in the switching decision by simulating the ACELP and TCXencoder and switch to the better performing branch. For this, the SNR ofthe ACELP and TCX branch are estimated based on an ACELP and TCXencoder/decoder simulation. The TCX encoder/decoder simulation isperformed without TNS/TTS analysis, IGF encoder,quantization-loop/arithmetic coder, or without any TCX decoder, Instead,the TCX SNR is estimated using an estimation of the quantizer distortionin the shaped MDCT domain. The ACELP encoder/decoder simulation isperformed using only a simulation of the adaptive codebook andinnovative codebook. The ACELP SNR is simply estimated by computing thedistortion introduced by a LTP filter in the weighted signal domain(adaptive codebook) and scaling this distortion by a constant factor(innovative codebook). Thus, the complexity is greatly reduced comparedto an approach where TCX and ACELP encoding is executed in parallel. Thebranch with the higher SNR is chosen for the subsequent completeencoding run.

In case the TCX branch is chosen, a TCX decoder is run in each framewhich outputs a signal at the ACELP sampling rate. This is used toupdate the memories used for the ACELT encoding path (LPC residual, Memw0, Memory deemphasis), to enable instant switching from TCX to ACELP.The memory update is performed in each TCX path.

Alternatively, a full analysis by synthesis process can performed, i.e.,both encoder simulators 621, 622 implement the actual encodingoperations and the results are compared by the selector 623.Alternatively, again, a complete feed forward calculation can be done byperforming a signal analysis. For example, when it is determined thatthe signal is a speech signal by a signal classifier the time domainencoder is selected and when it is determined that the signal is a musicsignal then the frequency domain encoder is selected. Other proceduresin order to distinguish between both encoders based on a signal analysisof the audio signal portion under consideration can also be applied.

Advantageously, the audio encoder additionally comprises across-processor 700 illustrated in FIG. 7A. When the frequency domainencoder 600 is active, the cross-processor 700 provides initializationdata to the time domain encoder 610 so that the time domain encoder isready for a seamless switch in a future signal portion. In other words,when the current signal portion is determined to be encoded using thefrequency domain encoder, and when it is determined by the controllerthat the immediately following audio signal portion is to be encoded bythe time domain encoder 610 then, without the cross-processor, such animmediate seamless switch would not be possible. The cross-processor,however, provides a signal derived from the frequency domain encoder 600to the time domain encoder 610 for the purpose of initializing memoriesin the time domain encoder since the time domain encoder 610 has adependency of a current frame from the input or encoded signal of animmediately in time preceding frame.

Hence, the time domain encoder 610 is configured to be initialized bythe initialization data in order to encode an audio signal portionfollowing an earlier audio signal portion encoded by the frequencydomain encoder 600 in an efficient manner.

In particular, the cross-processor comprises a time converter forconverting a frequency domain representation into a time domainrepresentation which can be forwarded to the time domain encoderdirectly or after some further processing. This converter is illustratedin FIG. 14A as an IMDCT (inverse modified discrete cosine transform)block. This block 702, however, has a different transform size comparedto the time-frequency converter block 602 indicated in FIG. 14A block(modified discrete cosine transform block). As indicated in block 602,the time-frequency converter 602 operates at the input sampling rate andthe inverse modified discrete cosine transform 702 operates at the lowerACELP sampling rate.

The ratio of the time domain coder sampling rate or ACELP sampling rateand the frequency domain coder sampling rate or input sampling rate canbe calculated and is a downsampling factor DS illustrated in FIG. 7B.The block 602 has a large transform size and the IMDCT block 702 has asmall transform size. As illustrated in FIG. 7B, the IMDCT block 702therefore comprises a selector 726 for selecting the lower spectralportion of an input into the IMDCT block 702. The portion of thefull-band spectrum is defined by the downsampling factor DS. Forexample, when the lower sampling rate is 16 kHz and the input samplingrate is 32 kHz then the downsampling factor is 0.5 and, therefore, theselector 726 selects the lower half of the full-band spectrum. When thespectrum has, for example, 1024 MDCT lines then the selector selects thelower 512 MDCT lines.

This low frequency portion of the full-band spectrum is input into asmall size transform and foldout block 720, as illustrated in FIG. 7B.The transform size is also selected in accordance with the downsamplingfactor and is 50% of the transform size in block 602. As synthesiswindowing with a window with a small number of coefficients is thenperformed. The number of coefficients of the synthesis window is equalto the downsampling factor multiplied by the number of coefficients ofthe analysis window used by block 602. Finally, an overlap add operationis performed with a smaller number of operations per block and thenumber of operations per block is again the number of operations perblock in a full rate implementation MDCT multiplied by the downsamplingfactor.

Thus, a very efficient downsampling operation can be applied since thedownsampling is included in the IMDCT implementation. In this context,it is emphasized that the block 702 can be implemented by an IMDCT butcan also be implemented by any other transform or filterbankimplementation which can be suitably sized in the actual transformkernel and other transform related operations.

In a further embodiment illustrated in FIG. 14A, the time-frequencyconverter comprises additional functionalities in addition to theanalyzer. The analyzer 604 of FIG. 6 may comprise in the embodiment ofFIG. 14A a temporal noise shaping/temporal tile shaping analysis block604 a operating as discussed in the context of FIG. 2B block 222 for theTNS/TTS analysis block 604 a and illustrated with respect to FIG. 2B forthe tonal mask 226 which corresponds to the IGF encoder 604 b in FIG.14A.

Furthermore, the frequency domain encoder may comprise a noise shapingblock 606 a. The noise shaping block 606 a is controlled by quantizedLPC coefficients as generated by block 1010. The quantized LPCcoefficients used for noise shaping 606 a perform a spectral shaping ofthe high resolution spectral values or spectral lines directly encoded(rather than parametrically encoded) and the result of block 606 a issimilar to the spectrum of a signal subsequent to an LPC filtering stageoperating in the time domain such as an LPC analysis filtering block 704to be described later on. Furthermore, the result of the noise shapingblock 606 a is then quantized and entropy coded as indicated by block606 b. The result of block 606 b corresponds to the encoded first audiosignal portion or a frequency domain coded audio signal portion(together with other side information).

The cross-processor 700 comprises a spectral decoder for calculating adecoded version of the first encoded signal portion. In the embodimentof FIG. 14A, the spectral decoder 701 comprises an inverse noise shapingblock 703, a gap filling decoder 704, a TNS/TTS synthesis block 705 andthe IMDCT block 702 discussed before. These blocks undo the specificoperations performed by blocks 602 to 606 b. In particular, a noiseshaping block 703 undoes the noise shaping performed by block 606 abased on the quantized LPC coefficients 1010. The IGF decoder 704operates as discussed with respect to FIG. 2A, blocks 202 and 206 andthe TNS/TTS synthesis block 705 operates as discussed in the context ofblock 210 of FIG. 2A and the spectral decoder additionally comprises theIMDCT block 702. Furthermore, the cross processor 700 in FIG. 14Aadditionally or alternatively comprises a delay stage 707 for feeding adelayed version of the decoded version obtained by the spectral decoder701 in a de-emphasis stage 617 of the second encoding processor for thepurpose of initializing the de-emphasis stage 617.

Furthermore, the cross-processor 17 may comprise in addition oralternatively a weighted prediction coefficient analysis filtering stage708 for filtering the decoded version and for feeding a filtered decodedversion to a codebook determinator 613 indicated as “MMSE” in FIG. 14Aof the second encoding processor for initializing this block.Additionally or alternatively, the cross-processor comprises the LPCanalysis filtering stage for filtering the decoded version of the firstencoded signal portion output by the spectral decoder 700 to an adaptivecodebook stage 712 for initialization of the block 612. In addition, oralternatively, the cross-processor also comprises a pre-emphasis stage709 for performing a pre-emphasis processing to the decoded versionoutput by a spectral decoder 701 before the LPC filtering. Thepre-emphasis stage output can also be fed to a further delay stage 710for the purpose of initializing an LPC synthesis filtering block 616within the time domain encoder 610 for the purpose of initializing thisLPC analysis filtering block 611.

The time domain encoder processor 610 comprises, as illustrated in FIG.14A, a pre-emphasis operating on the lower ACELP sampling rate. Asillustrated, this pre-emphasis is the pre-emphasis performed in thepreprocessing stage 1000 and has reference number 1005. The pre-emphasisdata is input into an LPC analysis filtering stage 611 operating in thetime domain and this filter is controlled by the quantized LPCcoefficients 1010 obtained by the preprocessing stage 1000. As knownfrom AMR-WB+ or USAC or other CELP encoders, the residual signalgenerated by block 611 is provided to an adaptive codebook 612 and,furthermore, the adaptive codebook 612 is connected to an innovativecodebook stage 614 and the codebook data from the adaptive codebook 612and from the innovative codebook are input into the bitstreammultiplexor as illustrated.

Furthermore, an ACELP gains/coding stage 612 is provided in series tothe innovative codebook stage 614 and the result of this block is inputinto a codebook determinator 613 indicated as MMSE in FIG. 14A. Thisblock cooperates with the innovative codebook block 614. Furthermore,the time domain encoder additionally comprises a decoder portion havingan LPC synthesis filtering block 616, a de-emphasis block 617 and anadaptive bass post filter stage 618 for calculating parameters for anadaptive bass post filter which is, however, applied at thedecoder-side. Without any adaptive bass post filtering on the decoderside, blocks 616, 617, 618 would not be necessary for the time domainencoder 610.

As illustrated, several blocks of the time domain decoder depend onprevious signals and these blocks are the adaptive codebook block, thecodebook determinator 613, the LPC synthesis filtering block 616 and thede-emphasis block 617. These blocks are provided with data from thecross-processor derived from the frequency domain encoding processordata in order to initialize these blocks for the purpose of being readyfor an instant switch from the frequency domain encoder to the timedomain encoder. As can also be seen from FIG. 14A, any dependence onearlier data is not necessary for the frequency domain encoder.Therefore, the cross-processor 700 does not provide any memoryinitialization data from the time domain encoder to the frequency domainencoder. However, for other implementations of the frequency domainencoder, where dependencies from the past exist and where memoryinitialization data is necessitated, the cross-processor 700 isconfigured to operate in both directions.

An embodiment of an audio encoder therefore comprises the followingparts:

The audio decoder is described in the following: The waveform decoderpart consists of a full-band TCX decoder path with IGF both operating atthe input sampling rate of the codec. In parallel, an alternative ACELPdecoder path at lower sampling rate exists that is reinforced furtherdownstream by a TD-BWE.

For ACELP initialization when switching from TCX to ACELP, a cross path(consisting of a shared TCX decoder frontend but additionally providingoutput at the lower sampling rate and some post-processing) exists thatperforms the inventive ACELP initialization. Sharing the same samplingrate and filter order between TCX and ACELP in the LPCs allows for aneasier and more efficient ACELP initialization.

For visualizing the switching, two switches are sketched in 14B. Whilethe second switch downstream chooses between TCX/IGF or ACELP/TD-BWEoutput, the first switch either pre-updates the buffers in theresampling QMF stage downstream the ACELP path by the output of thecross path or simply passes on the ACELP output.

Subsequently, audio decoder implementations in accordance with aspectsof the present invention are discussed in the context of FIGS. 11A-14C.

An audio decoder for decoding an encoded audio signal 1101 comprises afirst decoding processor 1120 for decoding a first encoded audio signalportion in a frequency domain. The first decoding processor 1120comprises a spectral decoder 1122 for decoding first spectral regionswith a high spectral resolution and for synthesizing second spectralregions using a parametric representation of the second spectral regionsand at least a decoded first spectral region to obtain a decodedspectral representation. The decoded spectral representation is afull-band decoded spectral representation as discussed in the context ofFIG. 6 and as also discussed in the context of FIG. 1A. Generally, thefirst decoding processor, therefore, comprises a full-bandimplementation with a gap filling procedure in the frequency domain. Thefirst decoding processor 1120 furthermore comprises a frequency-timeconverter 1124 for converting the decoded spectral representation into atime domain to obtain a decoded first audio signal portion.

Furthermore, the audio decoder comprises a second decoding processor1140 for decoding the second encoded audio signal portion in the timedomain to obtain a decoded second signal portion. Furthermore, the audiodecoder comprises a combiner 1160 for combining the decoded first signalportion and the decoded second signal portion to obtain a decoded audiosignal. The decoded signal portions are combined in sequence which isalso illustrated in FIG. 14B by a switch implementation 1160representing an embodiment of the combiner 1160 of FIG. 11A.

Advantageously, the second decoding processor 1140 is a time domainbandwidth extension processor and comprises, as illustrated in FIG. 12 ,a time domain low band decoder 1200 for decoding a low band time domainsignal. This implementation furthermore comprises an upsampler 1210 forupsampling the low band time domain signal. Additionally, a time domainbandwidth extension decoder 1220 is provided for synthesizing a highband of the output audio signal. Furthermore, a mixer 1230 is providedfor mixing a synthesized high band of the time domain output signal andan upsampled low band time domain signal to obtain the time domainencoder output. Hence, block 1140 in FIG. 11A can be implemented by thefunctionality of FIG. 12 in an embodiment.

FIG. 13 illustrates an embodiment of the time domain bandwidth extensiondecoder 1220 of FIG. 12 . Advantageously, a time domain upsampler 1221is provided which receives, as an input, an LPC residual signal from atime domain low band decoder included within block 1140 and illustratedat 1200 in FIG. 12 and further illustrated in the context of FIG. 14B.The time domain upsampler 1221 generates an upsampled version of the LPCresidual signal. This version is then input into a non-linear distortionblock 1222 which generates, based on its input signal, an output signalhaving higher frequency values. A non-linear distortion can be acopy-up, a mirroring, a frequency shift or a non-linear device such as adiode or a transistor operated in the non-linear region. The outputsignal of block 1222 is input into an LPC synthesis filtering block 1223which is controlled by LPC data used for the low band decoder as well orby specific envelope data generated by the time domain bandwidthextension block 920 on the encoder-side of FIG. 14A, for example. Theoutput of the LPC synthesis block is then input into a bandpass orhighpass filter 1224 to finally obtain the high band, which is theninput into the mixer 1230 as illustrated in FIG. 12 .

Subsequently, an implementation of the upsampler 1210 of FIG. 12 isdiscussed in the context of FIG. 14B. The upsampler may comprise ananalysis filterbank operating at a first time domain low band decodersampling rate. A specific implementation of such an analysis filterbankis a QMF analysis filterbank 1471 illustrated in FIG. 14B. Furthermore,the upsampler comprises a synthesis filterbank 1473 operating at asecond output sampling rate being higher than the first time domain lowband sampling rate. Hence, the QMF synthesis filterbank 1473 which is animplementation of the general filterbank operates at the output samplingrate. When the downsampling factor T as discussed in the context of FIG.7B is 0.5, then the QMF analysis filterbank 1471 has, e.g. only 32filterbank channels and the QMF synthesis filterbank 1473 has e.g. 64QMF channels, but the higher half of the filterbank channels, i.e., theupper 32 filterbank channels are fed with zeroes or noise, while thelower 32 filterbank channels are fed with the corresponding signalsprovided by the QMF analysis filterbank 1471. Advantageously, however, abandpass filtering 1472 is performed within the QMF filterbank domain inorder to make sure that the QMF synthesis output 1473 is an upsampledversion of the ACELP decoder output, but without any artifacts above themaximum frequency of the ACELP decoder.

Further processing operations can be performed within the QMF domain inaddition or instead of the bandpass filtering 1472. If no processing isperformed at all, then the QMF analysis and the QMF synthesis constitutean efficient upsampler 1210.

Subsequently, the construction of the individual elements in FIG. 14Bare discussed in more detail.

The full-band frequency domain decoder 1120 comprises a first decodingblock 1122 a for decoding the high resolution spectral coefficients andfor additionally performing noise filling in the low band portion asknown, for example, from the USAC technology. Furthermore, the full-banddecoder comprises an IGF processor 1122 b for filling the spectral holesusing synthesized spectral values which have been only parametricallyand, therefore, encoded with a low resolution on the encoder-side. Then,in block 1122 c, an inverse noise shaping is performed and the result isinput into a TNS/TTS synthesis block 705 which provides, as a finaloutput, an input to a frequency-time converter 1124, which may beimplemented as an inverse modified discrete cosine transform operatingat the output, i.e., high sampling rate.

Furthermore, a harmonic or LTP post-filter is used which is controlledby data obtained by the TCX LTP parameter extraction block 1006 in FIG.14B. The result is then the decoded first audio signal portion at theoutput sampling rate and as can be seen from FIG. 14B, this data has thehigh sampling rate and, therefore, any further frequency enhancement isnot necessary at all due to the fact that the decoding processor is afrequency domain full-band decoder advantageously operating using theintelligent gap filling technology discussed in the context of FIGS.1A-5C.

Several elements in FIG. 14B are quite similar to the correspondingblocks in the cross-processor 700 of FIG. 14A, particularly with respectto the IGF decoder 704 corresponding to IGF processing 1122 b and theinverse noise shaping operation controlled by quantized LPC coefficients1145 corresponds to the inverse noise shaping 703 of FIG. 14A and theTNS/TTS synthesis block 705 in FIG. 14B corresponds to the block TNS/TTSsynthesis 705 in FIG. 14A. Importantly, however, the IMDCT block 1124 inFIG. 14B operates at the high sampling rate while the IMDCT block 702 inFIG. 14A operates at a low sampling rate. Hence, the block 1124 in FIG.14B comprises the large sized transform and fold-out block 710, thesynthesis window in block 712 and the overlap-add stage 714 with thecorresponding large number of operations, large number of windowcoefficients and a large transform size compared to the correspondingfeatures 720, 722, 724, which are operated in block 702, and as will beoutlined later on, in block 1171 of the cross-processor 1170 in FIG. 14Bas well.

The time domain decoding processor 1140 may comprise the ACELP or timedomain low band decoder 1200 comprising an ACELP decoder stage 1149 forobtaining decoded gains and the innovative codebook information.Additionally, an ACELP adaptive codebook stage 1141 is provided and asubsequent ACELP post-processing stage 1142 and a final synthesis filtersuch as LPC synthesis filter 1143, which is again controlled by thequantized LPC coefficients 1145 obtained from the bitstreamdemultiplexer 1100 corresponding to the encoded signal parser 1100 inFIG. 11A. The output of the LPC synthesis filter 1143 is input into ade-emphasis stage 1144 for canceling or undoing the processingintroduced by the pre-emphasis stage 1005 of the pre-processor 1000 ofFIG. 14A. The result is the time domain output signal at a low samplingrate and a low band and in case the frequency domain output isnecessitated, the switch 1480 is in the indicated position and theoutput of the de-emphasis stage 1144 is introduced into the upsampler1210 and then mixed with the high bands from the time domain bandwidthextension decoder 1220.

In accordance with embodiments of the present invention, the audiodecoder additionally comprises the cross-processor 1170 illustrated inFIG. 11B and in FIG. 14B for calculating, from the decoded spectralrepresentation of the first encoded audio signal portion, initializationdata of the second decoding processor so that the second decodingprocessor is initialized to decode the encoded second audio signalportion following in time the first audio signal portion in the encodedaudio signal, i.e., such that the time domain decoding processor 1140 isready for an instant switch from one audio signal portion to the nextwithout any loss in quality or efficiency.

Advantageously, the cross-processor 1170 comprises an additionalfrequency-time converter 1171 operating at a lower sampling rate thanthe frequency-time converter of the first decoding processor in order toobtain a further decoded first signal portion in the time domain to beused as the initialization signal or for which any initialization datacan be derived. Advantageously, this IMDCT or low sampling ratefrequency-time converter is implemented as illustrated in FIG. 7B, item726 (selector), item 720 (small-size transform and fold-out), synthesiswindowing with a smaller number of window coefficients as indicated in722 and an overlap-add stage with a smaller number of operations asindicated at 724. Hence, the IMDCT block 1124 in the frequency domainfull-band decoder is implemented as indicated by block 710, 712, 714,and the IMDCT block 1171 is implemented as indicated in FIG. 7B by block726, 720, 722, 724. Again, the downsampling factor is the ratio betweenthe time domain coder sampling rate or the low sampling rate and thehigher frequency domain sampling rate or output sampling rate and thisdownsampling factor is lower than 1 and can be any number greater than 0and lower than 1.

As illustrated in FIG. 14B, the cross-processor 1170 further comprises,alone or in addition to other elements, a delay stage 1172 for delayingthe further decoded first signal portion and for feeding the delayeddecoded first signal portion into a de-emphasis stage 1144 of the seconddecoding processor for initialization. Furthermore, the cross-processorcomprises, in addition or alternatively, a pre-emphasis filter 1173 anda delay stage 1175 for filtering and delaying a further decoded firstsignal portion and for providing the delayed output of block 1175 intoan LPC synthesis filtering stage 1143 of the ACELP decoder for thepurpose of initialization.

Furthermore, the cross-processor may comprise alternatively or inaddition to the other mentioned elements an LPC analysis filter 1174 forgenerating a prediction residual signal from the further decoded firstsignal portion or a pre-emphasized further decoded first signal portionand for feeding the data into a codebook synthesizer of the seconddecoding processor and advantageously, into the adaptive codebook stage1141. Furthermore, the output of the frequency-time converter 1171 withthe low sampling rate is also input into the QMF analysis stage 1471 ofthe upsampler 1210 for the purpose of initialization, i.e., when thecurrently decoded audio signal portion is delivered by the frequencydomain full-band decoder 1120.

The audio decoder is described in the following: The waveform decoderpart consists of a full-band TCX decoder path with IGF both operating atthe input sampling rate of the codec. In parallel, an alternative ACELPdecoder path at lower sampling rate exists that is reinforced furtherdownstream by a TD-BWE.

For ACELP initialization when switching from TCX to ACELP, a cross path(consisting of a shared TCX decoder frontend but additionally providingoutput at the lower sampling rate and some post-processing) exists thatperforms the inventive ACELP initialization. Sharing the same samplingrate and filter order between TCX and ACELP in the LPCs allows for aneasier and more efficient ACELP initialization.

For visualizing the switching, two switches are sketched in FIG. 14B.While the second switch downstream chooses between TCX/IGF orACELP/TD-BWE output, the first switch either pre-updates the buffers inthe resampling QMF stage downstream the ACELP path by the output of thecross path or simply passes on the ACELP output.

To summarize, advantageous aspects of the invention which can be usedalone or in combination relate to a combination of an ACELP and TD-BWEcoder with a full-band capable TCX/IGF technology advantageouslyassociated with using a cross signal.

A further specific feature is a cross signal path for the ACELPinitialization to enable seamless switching.

A further aspect is that a short IMDCT is fed with a lower part ofhigh-rate long MDCT coefficients to efficiently implement a sample rateconversion in the cross-path.

A further feature is an efficient realization of the cross-path partlyshared with a full-band TCX/IGF in the decoder.

A further feature is the cross signal path for the QMF initialization toenable seamless switching from TCX to ACELP.

An additional feature is a cross-signal path to the QMF allowingcompensating the delay gap between ACELP resampled output and afilterbank-TCX/IGF output when switching from ACELP to TCX.

A further aspect is that an LPC is provided for both the TCX and theACELP coder at the same sampling rate and filter order, although theTCX/IGF encoder/decoder is full-band capable.

Subsequently, FIG. 14C is discussed as an implementation of a timedomain decoder operating either as a stand-alone decoder or in thecombination with the full-band capable frequency domain decoder.

Generally, the time domain decoder comprises an ACELP decoder, asubsequently connected resampler or upsampler and a time domainbandwidth extension functionality. Particularly, the ACELP decodercomprises an ACELP decoding stage for restoring gains and the innovativecodebook 1149, an ACELP-adaptive codebook stage 1141, an ACELPpost-processor 1142, an LPC synthesis filter 1143 controlled byquantized LPC coefficients from a bitstream demultiplexer or encodedsignal parser and the subsequently connected de-emphasis stage 1144.Advantageously, the time domain residual signal being at an ACELPsampling rate is input into a time domain bandwidth extension decoder1220 which provides a high band at the outputs.

In order to upsample the de-emphasis 1144 output, an upsamplercomprising the QMF analysis block 1471, and the QMF synthesis block 1473are provided. Within the filterbank domain defined by blocks 1471 and1473, a bandpass filter may be applied. Particularly, as has beendiscussed before, the same functionalities can also be used which havebeen discussed with respect to the same reference numbers. Furthermore,the time domain bandwidth extension decoder 1220 can be implemented asillustrated in FIG. 13 and, generally, comprises an upsampling of theACELP residual signal or time domain residual signal at the ACELPsampling rate finally to an output sampling rate of the bandwidthextended signal.

Subsequently, further details with respect to the frequency domainencoder and decoder being full-band capable are discussed with respectto FIGS. 1A-5C.

FIG. 1A illustrates an apparatus for encoding an audio signal 99. Theaudio signal 99 is input into a time spectrum converter 100 forconverting an audio signal having a sampling rate into a spectralrepresentation 101 output by the time spectrum converter. The spectrum101 is input into a spectral analyzer 102 for analyzing the spectralrepresentation 101. The spectral analyzer 101 is configured fordetermining a first set of first spectral portions 103 to be encodedwith a first spectral resolution and a different second set of secondspectral portions 105 to be encoded with a second spectral resolution.The second spectral resolution is smaller than the first spectralresolution. The second set of second spectral portions 105 is input intoa parameter calculator or parametric coder 104 for calculating spectralenvelope information having the second spectral resolution. Furthermore,a spectral domain audio coder 106 is provided for generating a firstencoded representation 107 of the first set of first spectral portionshaving the first spectral resolution. Furthermore, the parametercalculator/parametric coder 104 is configured for generating a secondencoded representation 109 of the second set of second spectralportions. The first encoded representation 107 and the second encodedrepresentation 109 are input into a bit stream multiplexer or bit streamformer 108 and block 108 finally outputs the encoded audio signal fortransmission or storage on a storage device.

Typically, a first spectral portion such as 306 of FIG. 3A will besurrounded by two second spectral portions such as 307 a, 307 b. This isnot the case in HE AAC, where the core coder frequency range is bandlimited

FIG. 1B illustrates a decoder matching with the encoder of FIG. 1A. Thefirst encoded representation 107 is input into a spectral domain audiodecoder 112 for generating a first decoded representation of a first setof first spectral portions, the decoded representation having a firstspectral resolution. Furthermore, the second encoded representation 109is input into a parametric decoder 114 for generating a second decodedrepresentation of a second set of second spectral portions having asecond spectral resolution being lower than the first spectralresolution.

The decoder further comprises a frequency regenerator 116 forregenerating a reconstructed second spectral portion having the firstspectral resolution using a first spectral portion. The frequencyregenerator 116 performs a tile filling operation, i.e., uses a tile orportion of the first set of first spectral portions and copies thisfirst set of first spectral portions into the reconstruction range orreconstruction band having the second spectral portion and typicallyperforms spectral envelope shaping or another operation as indicated bythe decoded second representation output by the parametric decoder 114,i.e., by using the information on the second set of second spectralportions. The decoded first set of first spectral portions and thereconstructed second set of spectral portions as indicated at the outputof the frequency regenerator 116 on line 117 is input into aspectrum-time converter 118 configured for converting the first decodedrepresentation and the reconstructed second spectral portion into a timerepresentation 119, the time representation having a certain highsampling rate.

FIG. 2B illustrates an implementation of the FIG. 1A encoder. An audioinput signal 99 is input into an analysis filterbank 220 correspondingto the time spectrum converter 100 of FIG. 1A. Then, a temporal noiseshaping operation is performed in TNS block 222. Therefore, the inputinto the spectral analyzer 102 of FIG. 1A corresponding to a block tonalmask 226 of FIG. 2B can either be full spectral values, when thetemporal noise shaping/temporal tile shaping operation is not applied orcan be spectral residual values, when the TNS operation as illustratedin FIG. 2B, block 222 is applied. For two-channel signals ormulti-channel signals, a joint channel coding 228 can additionally beperformed, so that the spectral domain encoder 106 of FIG. 1A maycomprise the joint channel coding block 228. Furthermore, an entropycoder 232 for performing a lossless data compression is provided whichis also a portion of the spectral domain encoder 106 of FIG. 1A.

The spectral analyzer/tonal mask 226 separates the output of TNS block222 into the core band and the tonal components corresponding to thefirst set of first spectral portions 103 and the residual componentscorresponding to the second set of second spectral portions 105 of FIG.1A. The block 224 indicated as IGF parameter extraction encodingcorresponds to the parametric coder 104 of FIG. 1A and the bitstreammultiplexer 230 corresponds to the bitstream multiplexer 108 of FIG. 1A.

Advantageously, the analysis filterbank 222 is implemented as an MDCT(modified discrete cosine transform filterbank) and the MDCT is used totransform the signal 99 into a time-frequency domain with the modifieddiscrete cosine transform acting as the frequency analysis tool.

The spectral analyzer 226 may apply a tonality mask. This tonality maskestimation stage is used to separate tonal components from thenoise-like components in the signal. This allows the core coder 228 tocode all tonal components with a psycho-acoustic module. The tonalitymask estimation stage can be implemented in numerous different ways andmay be implemented similar in its functionality to the sinusoidal trackestimation stage used in sine and noise-modeling for speech/audio coding[8, 9] or an HILN model based audio coder described in [10].Advantageously, an implementation is used which is easy to implementwithout the need to maintain birth-death trajectories, but any othertonality or noise detector can be used as well.

The IGF module calculates the similarity that exists between a sourceregion and a target region. The target region will be represented by thespectrum from the source region. The measure of similarity between thesource and target regions is done using a cross-correlation approach.The target region is split into nTar non-overlapping frequency tiles.For every tile in the target region, nSrc source tiles are created froma fixed start frequency. These source tiles overlap by a factor between0 and 1, where 0 means 0% overlap and 1 means 100% overlap. Each ofthese source tiles is correlated with the target tile at various lags tofind the source tile that best matches the target tile. The bestmatching tile number is stored in tileNum[idx_tar], the lag at which itbest correlates with the target is stored in xcorr_lag[idx_tar][idx_src]and the sign of the correlation is stored inxcorr_sign[ids_tar][idx_src]. In case the correlation is highlynegative, the source tile needs to be multiplied by −1 before the tilefilling process at the decoder. The IGF module also takes care of notoverwriting the tonal components in the spectrum since the tonalcomponents are preserved using the tonality mask. A band-wise energyparameter is used to store the energy of the target region enabling usto reconstruct the spectrum accurately.

This method has certain advantages over the classical SBR [1] in thatthe harmonic grid of a multi-tone signal is preserved by the core coderwhile only the gaps between the sinusoids is filled with the bestmatching “shaped noise” from the source region. Another advantage ofthis system compared to ASR (Accurate Spectral Replacement) [2-4] is theabsence of a signal synthesis stage which creates the important portionsof the signal at the decoder. Instead, this task is taken over by thecore coder, enabling the preservation of important components of thespectrum. Another advantage of the proposed system is the continuousscalability that the features offer. Just using tileNum[idx_tar] andxcorr_lag=0, for every tile is called gross granularity matching and canbe used for low bitrates while using variable xcorr_lag for every tileenables us to match the target and source spectra better.

In addition, a tile choice stabilization technique is proposed whichremoves frequency domain artifacts such as trilling and musical noise.

In case of stereo channel pairs an additional joint stereo processing isapplied. This is done, because for a certain destination range thesignal can a highly correlated panned sound source. In case the sourceregions chosen for this particular region are not well correlated,although the energies are matched for the destination regions, thespatial image can suffer due to the uncorrelated source regions. Theencoder analyses each destination region energy band, typicallyperforming a cross-correlation of the spectral values and if a certainthreshold is exceeded, sets a joint flag for this energy band. In thedecoder the left and right channel energy bands are treated individuallyif this joint stereo flag is not set. In case the joint stereo flag isset, both the energies and the patching are performed in the jointstereo domain. The joint stereo information for the IGF regions issignaled similar the joint stereo information for the core coding,including a flag indicating in case of prediction if the direction ofthe prediction is from downmix to residual or vice versa.

The energies can be calculated from the transmitted energies in theL/R-domain.

midNrg[k]=leftNrg[k]+rightNrg[k];

sidNrg[k]=leftNrg[k]−rightNrg[k];

With k being the frequency index in the transform domain.

Another solution is to calculate and transmit the energies directly inthe joint stereo domain for bands where joint stereo is active, so noadditional energy transformation is needed at the decoder side.

The source tiles are created according to the Mid/Side-Matrix:

midTile[k]=0.5·(leftTile[k]+rightTile[k])

sideTile [k]=0.5·(leftTile[k]−rightTile[k])

Energy adjustment:

midTile[k]=midTile[k]*midNrg[k];

sideTile[k]=sideTile[k]*sideNrg[k];

Joint stereo->LR transformation:

If no additional prediction parameter is coded:

leftTile[k]=midTile[k]+sideTile[k]

rightTile[k]=midTile[k]−sideTile[k]

If an additional prediction parameter is coded and if the signalleddirection is from mid to side:

sideTile[k]=sideTile[k]−predictionCoeff·midTile[k]

leftTile[k]=midTile[k]+sideTile[k]

rightTile[k]=midTile[k]−sideTile[k]

If the signalled direction is from side to mid:

midTile1[k]=midTile[k]−predictionCoeff·sideTile[k]

leftTile[k]=midTile1[k]−sideTile[k]

rightTile[k]=midTile1[k]+sideTile[k]

This processing ensures that from the tiles used for regenerating highlycorrelated destination regions and panned destination regions, theresulting left and right channels still represent a correlated andpanned sound source even if the source regions are not correlated,preserving the stereo image for such regions.

In other words, in the bitstream, joint stereo flags are transmittedthat indicate whether L/R or M/S as an example for the general jointstereo coding shall be used. In the decoder, first, the core signal isdecoded as indicated by the joint stereo flags for the core bands.Second, the core signal is stored in both L/R and M/S representation.For the IGF tile filling, the source tile representation is chosen tofit the target tile representation as indicated by the joint stereoinformation for the IGF bands.

Temporal Noise Shaping (TNS) is a standard technique and part of AAC[11-13]. TNS can be considered as an extension of the basic scheme of aperceptual coder, inserting an optional processing step between thefilterbank and the quantization stage. The main task of the TNS moduleis to hide the produced quantization noise in the temporal maskingregion of transient like signals and thus it leads to a more efficientcoding scheme. First, TNS calculates a set of prediction coefficientsusing “forward prediction” in the transform domain, e.g. MDCT. Thesecoefficients are then used for flattening the temporal envelope of thesignal. As the quantization affects the TNS filtered spectrum, also thequantization noise is temporarily flat. By applying the invers TNSfiltering on decoder side, the quantization noise is shaped according tothe temporal envelope of the TNS filter and therefore the quantizationnoise gets masked by the transient.

IGF is based on an MDCT representation. For efficient coding,advantageously long blocks of approx. 20 ms have to be used. If thesignal within such a long block contains transients, audible pre- andpost-echoes occur in the IGF spectral bands due to the tile filling.FIG. 7C shows a typical pre-echo effect before the transient onset dueto IGF. On the left side, the spectrogram of the original signal isshown and on the right side the spectrogram of the bandwidth extendedsignal without TNS filtering is shown.

This pre-echo effect is reduced by using TNS in the IGF context. Here,TNS is used as a temporal tile shaping (TTS) tool as the spectralregeneration in the decoder is performed on the TNS residual signal. Thenecessitated TTS prediction coefficients are calculated and appliedusing the full spectrum on encoder side as usual. The TNS/TTS start andstop frequencies are not affected by the IGF start frequency figfstartof the IGF tool. In comparison to the legacy TNS, the TTS stop frequencyis increased to the stop frequency of the IGF tool, which is higherfigfstart. On decoder side the TNS/TTS coefficients are applied on thefull spectrum again, i.e. the core spectrum plus the regeneratedspectrum plus the tonal components from the tonality map (see FIG. 7E).The application of TTS is done to form the temporal envelope of theregenerated spectrum to match the envelope of the original signal again.So the shown pre-echoes are reduced. In addition, it still shapes thequantization noise in the signal below figfstart as usual with TNS.

In legacy decoders, spectral patching on an audio signal corruptsspectral correlation at the patch borders and thereby impairs thetemporal envelope of the audio signal by introducing dispersion. Hence,another benefit of performing the IGF tile filling on the residualsignal is that, after application of the shaping filter, tile bordersare seamlessly correlated, resulting in a more faithful temporalreproduction of the signal.

In an inventive encoder, the spectrum having undergone TNS/TTSfiltering, tonality mask processing and IGF parameter estimation isdevoid of any signal above the IGF start frequency except for tonalcomponents. This sparse spectrum is now coded by the core coder usingprinciples of arithmetic coding and predictive coding. These codedcomponents along with the signaling bits form the bitstream of theaudio.

FIG. 2A illustrates the corresponding decoder implementation. Thebitstream in FIG. 2A corresponding to the encoded audio signal is inputinto the demultiplexer/decoder which would be connected, with respect toFIG. 1B, to the blocks 112 and 114. The bitstream demultiplexerseparates the input audio signal into the first encoded representation107 of FIG. 1B and the second encoded representation 109 of FIG. 1B. Thefirst encoded representation having the first set of first spectralportions is input into the joint channel decoding block 204corresponding to the spectral domain decoder 112 of FIG. 1B. The secondencoded representation is input into the parametric decoder 114 notillustrated in FIG. 2A and then input into the IGF block 202corresponding to the frequency regenerator 116 of FIG. 1B. The first setof first spectral portions necessitated for frequency regeneration areinput into IGF block 202 via line 203. Furthermore, subsequent to jointchannel decoding 204 the specific core decoding is applied in the tonalmask block 206 so that the output of tonal mask 206 corresponds to theoutput of the spectral domain decoder 112. Then, a combination bycombiner 208 is performed, i.e., a frame building where the output ofcombiner 208 now has the full range spectrum, but still in the TNS/TTSfiltered domain. Then, in block 210, an inverse TNS/TTS operation isperformed using TNS/TTS filter information provided via line 109, i.e.,the TTS side information may be included in the first encodedrepresentation generated by the spectral domain encoder 106 which can,for example, be a straightforward AAC or USAC core encoder, or can alsobe included in the second encoded representation. At the output of block210, a complete spectrum until the maximum frequency is provided whichis the full range frequency defined by the sampling rate of the originalinput signal. Then, a spectrum/time conversion is performed in thesynthesis filterbank 212 to finally obtain the audio output signal.

FIG. 3A illustrates a schematic representation of the spectrum. Thespectrum is subdivided in scale factor bands SCB where there are sevenscale factor bands SCB1 to SCB7 in the illustrated example of FIG. 3A.The scale factor bands can be AAC scale factor bands which are definedin the AAC standard and have an increasing bandwidth to upperfrequencies as illustrated in FIG. 3A schematically. It is of advantageto perform intelligent gap filling not from the very beginning of thespectrum, i.e., at low frequencies, but to start the IGF operation at anIGF start frequency illustrated at 309. Therefore, the core frequencyband extends from the lowest frequency to the IGF start frequency. Abovethe IGF start frequency, the spectrum analysis is applied to separatehigh resolution spectral components 304, 305, 306, 307 (the first set offirst spectral portions) from low resolution components represented bythe second set of second spectral portions. FIG. 3A illustrates aspectrum which is exemplarily input into the spectral domain encoder 106or the joint channel coder 228, i.e., the core encoder operates in thefull range, but encodes a significant amount of zero spectral values,i.e., these zero spectral values are quantized to zero or are set tozero before quantizing or subsequent to quantizing. Anyway, the coreencoder operates in full range, i.e., as if the spectrum would be asillustrated, i.e., the core decoder does not necessarily have to beaware of any intelligent gap filling or encoding of the second set ofsecond spectral portions with a lower spectral resolution.

Advantageously, the high resolution is defined by a line-wise coding ofspectral lines such as MDCT lines, while the second resolution or lowresolution is defined by, for example, calculating only a singlespectral value per scale factor band, where a scale factor band coversseveral frequency lines. Thus, the second low resolution is, withrespect to its spectral resolution, much lower than the first or highresolution defined by the line-wise coding typically applied by the coreencoder such as an AAC or USAC core encoder.

Regarding scale factor or energy calculation, the situation isillustrated in FIG. 3B. Due to the fact that the encoder is a coreencoder and due to the fact that there can, but does not necessarilyhave to be, components of the first set of spectral portions in eachband, the core encoder calculates a scale factor for each band not onlyin the core range below the IGF start frequency 309, but also above theIGF start frequency until the maximum frequency figfstop which issmaller or equal to the half of the sampling frequency, i.e., f_(s/2).Thus, the encoded tonal portions 302, 304, 305, 306, 307 of FIG. 3A and,in this embodiment together with the scale factors SCB1 to SCB7correspond to the high resolution spectral data. The low resolutionspectral data are calculated starting from the IGF start frequency andcorrespond to the energy information values E₁, E₂, E₃, E₄, which aretransmitted together with the scale factors SF4 to SF7.

Particularly, when the core encoder is under a low bitrate condition, anadditional noise-filling operation in the core band, i.e., lower infrequency than the IGF start frequency, i.e., in scale factor bands SCB1to SCB3 can be applied in addition. In noise-filling, there existseveral adjacent spectral lines which have been quantized to zero. Onthe decoder-side, these quantized to zero spectral values arere-synthesized and the re-synthesized spectral values are adjusted intheir magnitude using a noise-filling energy such as NF₂ illustrated at308 in FIG. 3B. The noise-filling energy, which can be given in absoluteterms or in relative terms particularly with respect to the scale factoras in USAC corresponds to the energy of the set of spectral valuesquantized to zero. These noise-filling spectral lines can also beconsidered to be a third set of third spectral portions which areregenerated by straightforward noise-filling synthesis without any IGFoperation relying on frequency regeneration using frequency tiles fromother frequencies for reconstructing frequency tiles using spectralvalues from a source range and the energy information E₁, E₂, E₃, E₄.

Advantageously, the bands, for which energy information is calculatedcoincide with the scale factor bands. In other embodiments, an energyinformation value grouping is applied so that, for example, for scalefactor bands 4 and 5, only a single energy information value istransmitted, but even in this embodiment, the borders of the groupedreconstruction bands coincide with borders of the scale factor bands. Ifdifferent band separations are applied, then certain re-calculations orsynchronization calculations may be applied, and this can make sensedepending on the certain implementation.

Advantageously, the spectral domain encoder 106 of FIG. 1A is apsycho-acoustically driven encoder as illustrated in FIG. 4A. Typically,as for example illustrated in the MPEG2/4 AAC standard or MPEG1/2, Layer3 standard, the to be encoded audio signal after having been transformedinto the spectral range (401 in FIG. 4A) is forwarded to a scale factorcalculator 400. The scale factor calculator is controlled by apsycho-acoustic model additionally receiving the to be quantized audiosignal or receiving, as in the MPEG1/2 Layer 3 or MPEG AAC standard, acomplex spectral representation of the audio signal. The psycho-acousticmodel calculates, for each scale factor band, a scale factorrepresenting the psycho-acoustic threshold. Additionally, the scalefactors are then, by cooperation of the well-known inner and outeriteration loops or by any other suitable encoding procedure adjusted sothat certain bitrate conditions are fulfilled. Then, the to be quantizedspectral values on the one hand and the calculated scale factors on theother hand are input into a quantizer processor 404. In thestraightforward audio encoder operation, the to be quantized spectralvalues are weighted by the scale factors and, the weighted spectralvalues are then input into a fixed quantizer typically having acompression functionality to upper amplitude ranges. Then, at the outputof the quantizer processor there do exist quantization indices which arethen forwarded into an entropy encoder typically having specific andvery efficient coding for a set of zero-quantization indices foradjacent frequency values or, as also called in the art, a “run” of zerovalues.

In the audio encoder of FIG. 1A, however, the quantizer processortypically receives information on the second spectral portions from thespectral analyzer. Thus, the quantizer processor 404 makes sure that, inthe output of the quantizer processor 404, the second spectral portionsas identified by the spectral analyzer 102 are zero or have arepresentation acknowledged by an encoder or a decoder as a zerorepresentation which can be very efficiently coded, specifically whenthere exist “runs” of zero values in the spectrum.

FIG. 4B illustrates an implementation of the quantizer processor. TheMDCT spectral values can be input into a set to zero block 410. Then,the second spectral portions are already set to zero before a weightingby the scale factors in block 412 is performed. In an additionalimplementation, block 410 is not provided, but the set to zerocooperation is performed in block 418 subsequent to the weighting block412. In an even further implementation, the set to zero operation canalso be performed in a set to zero block 422 subsequent to aquantization in the quantizer block 420. In this implementation, blocks410 and 418 would not be present. Generally, at least one of the blocks410, 418, 422 are provided depending on the specific implementation.

Then, at the output of block 422, a quantized spectrum is obtainedcorresponding to what is illustrated in FIG. 3A. This quantized spectrumis then input into an entropy coder such as 232 in FIG. 2B which can bea Huffman coder or an arithmetic coder as, for example, defined in theUSAC standard.

The set to zero blocks 410, 418, 422, which are provided alternativelyto each other or in parallel are controlled by the spectral analyzer424. The spectral analyzer may comprise any implementation of awell-known tonality detector or comprises any different kind of detectoroperative for separating a spectrum into components to be encoded with ahigh resolution and components to be encoded with a low resolution.Other such algorithms implemented in the spectral analyzer can be avoice activity detector, a noise detector, a speech detector or anyother detector deciding, depending on spectral information or associatedmetadata on the resolution requirements for different spectral portions.

FIG. 5A illustrates an implementation of the time spectrum converter 100of FIG. 1A as, for example, implemented in AAC or USAC. The timespectrum converter 100 comprises a windower 502 controlled by atransient detector 504. When the transient detector 504 detects atransient, then a switchover from long windows to short windows issignaled to the windower. The windower 502 then calculates, foroverlapping blocks, windowed frames, where each windowed frame typicallyhas two N values such as 2048 values. Then, a transformation within ablock transformer 506 is performed, and this block transformer typicallyadditionally provides a decimation, so that a combineddecimation/transform is performed to obtain a spectral frame with Nvalues such as MDCT spectral values. Thus, for a long window operation,the frame at the input of block 506 comprises two N values such as 2048values and a spectral frame then has 1024 values. Then, however, aswitch is performed to short blocks, when eight short blocks areperformed where each short block has ⅛ windowed time domain valuescompared to a long window and each spectral block has ⅛ spectral valuescompared to a long block. Thus, when this decimation is combined with a50% overlap operation of the windower, the spectrum is a criticallysampled version of the time domain audio signal 99.

Subsequently, reference is made to FIG. 5B illustrating a specificimplementation of frequency regenerator 116 and the spectrum-timeconverter 118 of FIG. 1B, or of the combined operation of blocks 208,212 of FIG. 2A. In FIG. 5B, a specific reconstruction band is consideredsuch as scale factor band 6 of FIG. 3A. The first spectral portion inthis reconstruction band, i.e., the first spectral portion 306 of FIG.3A is input into the frame builder/adjustor block 510. Furthermore, areconstructed second spectral portion for the scale factor band 6 isinput into the frame builder/adjuster 510 as well. Furthermore, energyinformation such as E₃ of FIG. 3B for a scale factor band 6 is alsoinput into block 510. The reconstructed second spectral portion in thereconstruction band has already been generated by frequency tile fillingusing a source range and the reconstruction band then corresponds to thetarget range. Now, an energy adjustment of the frame is performed tothen finally obtain the complete reconstructed frame having the N valuesas, for example, obtained at the output of combiner 208 of FIG. 2A.Then, in block 512, an inverse block transform/interpolation isperformed to obtain 248 time domain values for the for example 124spectral values at the input of block 512. Then, a synthesis windowingoperation is performed in block 514 which is again controlled by a longwindow/short window indication transmitted as side information in theencoded audio signal. Then, in block 516, an overlap/add operation witha previous time frame is performed. Advantageously, MDCT applies a 50%overlap so that, for each new time frame of 2N values, N time domainvalues are finally output. A 50% overlap is heavily of advantage due tothe fact that it provides critical sampling and a continuous crossoverfrom one frame to the next frame due to the overlap/add operation inblock 516.

As illustrated at 301 in FIG. 3A, a noise-filling operation canadditionally be applied not only below the IGF start frequency, but alsoabove the IGF start frequency such as for the contemplatedreconstruction band coinciding with scale factor band 6 of FIG. 3A.Then, noise-filling spectral values can also be input into the framebuilder/adjuster 510 and the adjustment of the noise-filling spectralvalues can also be applied within this block or the noise-fillingspectral values can already be adjusted using the noise-filling energybefore being input into the frame builder/adjuster 510.

Advantageously, an IGF operation, i.e., a frequency tile fillingoperation using spectral values from other portions can be applied inthe complete spectrum. Thus, a spectral tile filling operation can notonly be applied in the high band above an IGF start frequency but canalso be applied in the low band. Furthermore, the noise-filling withoutfrequency tile filling can also be applied not only below the IGF startfrequency but also above the IGF start frequency. It has, however, beenfound that high quality and high efficient audio encoding can beobtained when the noise-filling operation is limited to the frequencyrange below the IGF start frequency and when the frequency tile fillingoperation is restricted to the frequency range above the IGF startfrequency as illustrated in FIG. 3A.

Advantageously, the target tiles (TT) (having frequencies greater thanthe IGF start frequency) are bound to scale factor band borders of thefull rate coder. Source tiles (ST), from which information is taken,i.e., for frequencies lower than the IGF start frequency are not boundby scale factor band borders. The size of the ST should correspond tothe size of the associated TT. This is illustrated using the followingexample. TT[0] has a length of 10 MDCT Bins. This exactly corresponds tothe length of two subsequent SCBs (such as 4+6). Then, all possible STthat are to be correlated with TT[0], have a length of 10 bins, too. Asecond target tile TT[1] being adjacent to TT[0] has a length of 15 binsI (SCB having a length of 7+8). Then, the ST for that have a length of15 bins rather than 10 bins as for TT[0].

Should the case arise that one cannot find a TT for an ST with thelength of the target tile (when e.g. the length of TT is greater thanthe available source range), then a correlation is not calculated andthe source range is copied a number of times into this TT (the copyingis done one after the other so that a frequency line for the lowestfrequency of the second copy immediately follows—in frequency—thefrequency line for the highest frequency of the first copy), until thetarget tile TT is completely filled up.

Subsequently, reference is made to FIG. 5C illustrating a furtherembodiment of the frequency regenerator 116 of FIG. 1B or the IGF block202 of FIG. 2A. Block 522 is a frequency tile generator receiving, notonly a target band ID, but additionally receiving a source band ID.Exemplarily, it has been determined on the encoder-side that the scalefactor band 3 of FIG. 3A is very well suited for reconstructing scalefactor band 7. Thus, the source band ID would be 2 and the target bandID would be 7. Based on this information, the frequency tile generator522 applies a copy up or harmonic tile filling operation or any othertile filling operation to generate the raw second portion of spectralcomponents 523. The raw second portion of spectral components has afrequency resolution identical to the frequency resolution included inthe first set of first spectral portions.

Then, the first spectral portion of the reconstruction band such as 307of FIG. 3A is input into a frame builder 524 and the raw second portion523 is also input into the frame builder 524. Then, the reconstructedframe is adjusted by the adjuster 526 using a gain factor for thereconstruction band calculated by the gain factor calculator 528.Importantly, however, the first spectral portion in the frame is notinfluenced by the adjuster 526, but only the raw second portion for thereconstruction frame is influenced by the adjuster 526. To this end, thegain factor calculator 528 analyzes the source band or the raw secondportion 523 and additionally analyzes the first spectral portion in thereconstruction band to finally find the correct gain factor 527 so thatthe energy of the adjusted frame output by the adjuster 526 has theenergy E₄ when a scale factor band 7 is contemplated.

In this context, it is very important to evaluate the high frequencyreconstruction accuracy of the present invention compared to HE-AAC.This is explained with respect to scale factor band 7 in FIG. 3A. It isassumed that a known encoder such as illustrated in FIG. 13A woulddetect the spectral portion 307 to be encoded with a high resolution asa “missing harmonics”. Then, the energy of this spectral component wouldbe transmitted together with a spectral envelope information for thereconstruction band such as scale factor band 7 to the decoder. Then,the decoder would recreate the missing harmonic. However, the spectralvalue, at which the missing harmonic 307 would be reconstructed by theknown decoder of FIG. 13B would be in the middle of band 7 at afrequency indicated by reconstruction frequency 390. Thus, the presentinvention avoids a frequency error 391 which would be introduced by theknown decoder of FIG. 13D.

In an implementation, the spectral analyzer is also implemented tocalculating similarities between first spectral portions and secondspectral portions and to determine, based on the calculatedsimilarities, for a second spectral portion in a reconstruction range afirst spectral portion matching with the second spectral portion as faras possible. Then, in this variable source range/destination rangeimplementation, the parametric coder will additionally introduce intothe second encoded representation a matching information indicating foreach destination range a matching source range. On the decoder-side,this information would then be used by a frequency tile generator 522 ofFIG. 5C illustrating a generation of a raw second portion 523 based on asource band ID and a target band ID.

Furthermore, as illustrated in FIG. 3A, the spectral analyzer isconfigured to analyze the spectral representation up to a maximumanalysis frequency being only a small amount below half of the samplingfrequency and advantageously being at least one quarter of the samplingfrequency or typically higher.

As illustrated, the encoder operates without downsampling and thedecoder operates without upsampling. In other words, the spectral domainaudio coder is configured to generate a spectral representation having aNyquist frequency defined by the sampling rate of the originally inputaudio signal.

Furthermore, as illustrated in FIG. 3A, the spectral analyzer isconfigured to analyze the spectral representation starting with a gapfilling start frequency and ending with a maximum frequency representedby a maximum frequency included in the spectral representation, whereina spectral portion extending from a minimum frequency up to the gapfilling start frequency belongs to the first set of spectral portionsand wherein a further spectral portion such as 304, 305, 306, 307 havingfrequency values above the gap filling frequency additionally isincluded in the first set of first spectral portions.

As outlined, the spectral domain audio decoder 112 is configured so thata maximum frequency represented by a spectral value in the first decodedrepresentation is equal to a maximum frequency included in the timerepresentation having the sampling rate wherein the spectral value forthe maximum frequency in the first set of first spectral portions iszero or different from zero. Anyway, for this maximum frequency in thefirst set of spectral components a scale factor for the scale factorband exists, which is generated and transmitted irrespective of whetherall spectral values in this scale factor band are set to zero or not asdiscussed in the context of FIGS. 3A and 3B.

The invention is, therefore, advantageous that with respect to otherparametric techniques to increase compression efficiency, e.g. noisesubstitution and noise filling (these techniques are exclusively forefficient representation of noise like local signal content) theinvention allows an accurate frequency reproduction of tonal components.To date, no state-of-the-art technique addresses the efficientparametric representation of arbitrary signal content by spectral gapfilling without the restriction of a fixed a-priory division in low band(LF) and high band (HF).

Embodiments of the inventive system improve the state-of-the-artapproaches and thereby provides high compression efficiency, no or onlya small perceptual annoyance and full audio bandwidth even for lowbitrates.

The general system consists of

-   -   full-band core coding    -   intelligent gap filling (tile filling or noise filling)    -   sparse tonal parts in core selected by tonal mask    -   joint stereo pair coding for full-band, including tile filling    -   TNS on tile    -   spectral whitening in IGF range

A first step towards a more efficient system is to remove the need fortransforming spectral data into a second transform domain different fromthe one of the core coder. As the majority of audio codecs, such as AACfor instance, use the MDCT as basic transform, it is useful to performthe BWE in the MDCT domain also. A second requirement for the BWE systemwould be the need to preserve the tonal grid whereby even HF tonalcomponents are preserved and the quality of the coded audio is thussuperior to the existing systems. To take care of both the abovementioned requirements for a BWE scheme, a new system is proposed calledIntelligent Gap Filling (IGF). FIG. 2B shows the block diagram of theproposed system on the encoder-side and FIG. 2A shows the system on thedecoder-side.

Subsequently, further optional features of the full band frequencydomain first encoding processor and the full band frequency domaindecoding processor incorporating the gap-filling operation, which can beimplemented separately or together are discussed and defined.

Particularly, the spectral domain decoder 112 corresponding to block1122 a is configured to output a sequence of decoded frames of spectralvalues, a decoded frame being the first decoded representation, whereinthe frame comprises spectral values for the first set of spectralportions and zero indications for the second spectral portions. Theapparatus for decoding furthermore comprises a combiner 208. Thespectral values are generated by a frequency regenerator for the secondset of second spectral portions, where both, the combiner and thefrequency regenerator are included within block 1122 b. Thus, bycombining the second spectral portions and the first spectral portions areconstructed spectral frame comprising spectral values for the firstset of the first spectral portions and the second set of spectralportions are obtained and the spectrum-time converter 118 correspondingto the IMDCT block 1124 in FIG. 14B then converts the reconstructedspectral frame into the time representation.

As outlined, the spectrum-time converter 118 or 1124 is configured toperform an inverse modified discrete cosine transform 512, 514 andfurther comprises an overlap-add stage 516 for overlapping and addingsubsequent time domain frames.

Particularly, the spectral domain audio decoder 1122 a is configured togenerate the first decoded representation so that the first decodedrepresentation has a Nyquist frequency defining a sampling rate beingequal to a sampling rate of the time representation generated by thespectrum-time converter 1124.

Furthermore, the decoder 1112 or 1122 a is configured to generate thefirst decoded representation so that a first spectral portion 306 isplaced with respect to frequency between two second spectral portions307 a, 307 b.

In a further embodiment, a maximum frequency represented by a spectralvalue for the maximum frequency in the first decoded representation isequal to a maximum frequency included in the time representationgenerated by the spectrum-time converter, wherein the spectral value forthe maximum frequency in the first representation is zero or differentfrom zero.

Furthermore, as illustrated in FIG. 3 the encoded first audio signalportion further comprises an encoded representation of a third set ofthird spectral portions to be reconstructed by noise filling, and thefirst decoding processor 1120 additionally includes a noise fillerincluded in block 1122 b for extracting noise filling information 308from an encoded representation of the third set of third spectralportions and for applying a noise filling operation in the third set ofthird spectral portions without using a first spectral portion in adifferent frequency range.

Furthermore, the spectral domain audio decoder 112 is configured togenerate the first decoded representation having the first spectralportions with the frequency values being greater than the frequencybeing equal to a frequency in the middle of the frequency range coveredby the time representation output by the spectrum-time converter 118 or1124.

Furthermore, the spectral analyzer or full-band analyzer 604 isconfigured to analyze the representation generated by the time-frequencyconverter 602 for determining a first set of first spectral portions tobe encoded with the first high spectral resolution and the differentsecond set of second spectral portions to be encoded with a secondspectral resolution which is lower than the first spectral resolutionand, by means of the spectral analyzer, a first spectral portion 306 isdetermined, with respect to frequency, between two second spectralportions in FIG. 3 at 307 a and 307 b.

Particularly, the spectral analyzer is configured for analyzing thespectral representation up to a maximum analysis frequency being atleast one quarter of a sampling frequency of the audio signal.

Particularly, the spectral domain audio encoder is configured to processa sequence of frames of spectral values for a quantization and entropycoding, wherein, in a frame, spectral values of the second set of secondportions are set to zero, or wherein, in the frame, spectral values ofthe first set of first spectral portions and the second set of thesecond spectral portions are present and wherein, during subsequentprocessing, spectral values in the second set of spectral portions areset to zero as exemplarily illustrated at 410, 418, 422.

The spectral domain audio encoder is configured to generate a spectralrepresentation having a Nyquist frequency defined by the sampling rateof the audio input signal or the first portion of the audio signalprocessed by the first encoding processor operating in the frequencydomain.

The spectral domain audio encoder 606 is furthermore configured toprovide the first encoded representation so that, for a frame of asampled audio signal, the encoded representation comprises the first setof first spectral portions and the second set of second spectralportions, wherein the spectral values in the second set of spectralportions are encoded as zero or noise values.

The full band analyzer 604 or 102 is configured to analyze the spectralrepresentation starting with the gap-filing start frequency 209 andending with a maximum frequency f_(max) represented by a maximumfrequency included in the spectral representation and a spectral portionextending from a minimum frequency up to the gap-filling start frequency309 belongs to the first set of first spectral portions.

Particularly, the analyzer is configured to apply a tonal maskprocessing at least of a portion of the spectral representation so thattonal components and non-tonal components are separated from each other,wherein the first set of the first spectral portions comprises the tonalcomponents and wherein the second set of the second spectral portionscomprises the non-tonal components.

Although the present invention has been described in the context ofblock diagrams where the blocks represent actual or logical hardwarecomponents, the present invention can also be implemented by acomputer-implemented method. In the latter case, the blocks representcorresponding method steps where these steps stand for thefunctionalities performed by corresponding logical or physical hardwareblocks.

Although some aspects have been described in the context of anapparatus, it is clear that these aspects also represent a descriptionof the corresponding method, where a block or device corresponds to amethod step or a feature of a method step. Analogously, aspectsdescribed in the context of a method step also represent a descriptionof a corresponding block or item or feature of a correspondingapparatus. Some or all of the method steps may be executed by (or using)a hardware apparatus, like for example, a microprocessor, a programmablecomputer or an electronic circuit. In some embodiments, some one or moreof the most important method steps may be executed by such an apparatus.

The inventive transmitted or encoded signal can be stored on a digitalstorage medium or can be transmitted on a transmission medium such as awireless transmission medium or a wired transmission medium such as theInternet.

Depending on certain implementation requirements, embodiments of theinvention can be implemented in hardware or in software. Theimplementation can be performed using a digital storage medium, forexample a floppy disc, a DVD, a Blu-Ray, a CD, a ROM, a PROM, and EPROM,an EEPROM or a FLASH memory, having electronically readable controlsignals stored thereon, which cooperate (or are capable of cooperating)with a programmable computer system such that the respective method isperformed. Therefore, the digital storage medium may be computerreadable.

Some embodiments according to the invention comprise a data carrierhaving electronically readable control signals, which are capable ofcooperating with a programmable computer system, such that one of themethods described herein is performed.

Generally, embodiments of the present invention can be implemented as acomputer program product with a program code, the program code beingoperative for performing one of the methods when the computer programproduct runs on a computer. The program code may, for example, be storedon a machine readable carrier.

Other embodiments comprise the computer program for performing one ofthe methods described herein, stored on a machine readable carrier.

In other words, an embodiment of the inventive method is, therefore, acomputer program having a program code for performing one of the methodsdescribed herein, when the computer program runs on a computer.

A further embodiment of the inventive method is, therefore, a datacarrier (or a non-transitory storage medium such as a digital storagemedium, or a computer-readable medium) comprising, recorded thereon, thecomputer program for performing one of the methods described herein. Thedata carrier, the digital storage medium or the recorded medium aretypically tangible and/or non-transitory.

A further embodiment of the invention method is, therefore, a datastream or a sequence of signals representing the computer program forperforming one of the methods described herein. The data stream or thesequence of signals may, for example, be configured to be transferredvia a data communication connection, for example, via the internet.

A further embodiment comprises a processing means, for example, acomputer or a programmable logic device, configured to, or adapted to,perform one of the methods described herein.

A further embodiment comprises a computer having installed thereon thecomputer program for performing one of the methods described herein.

A further embodiment according to the invention comprises an apparatusor a system configured to transfer (for example, electronically oroptically) a computer program for performing one of the methodsdescribed herein to a receiver. The receiver may, for example, be acomputer, a mobile device, a memory device or the like. The apparatus orsystem may, for example, comprise a file server for transferring thecomputer program to the receiver.

In some embodiments, a programmable logic device (for example, a fieldprogrammable gate array) may be used to perform some or all of thefunctionalities of the methods described herein. In some embodiments, afield programmable gate array may cooperate with a microprocessor inorder to perform one of the methods described herein. Generally, themethods may be performed by any hardware apparatus.

While this invention has been described in terms of several embodiments,there are alterations, permutations, and equivalents which will beapparent to others skilled in the art and which fall within the scope ofthis invention. It should also be noted that there are many alternativeways of implementing the methods and compositions of the presentinvention. It is therefore intended that the following appended claimsbe interpreted as including all such alterations, permutations, andequivalents as fall within the true spirit and scope of the presentinvention.

1. An audio encoder for encoding an audio signal, the audio signalcomprising a first audio signal portion and a timely subsequent secondaudio signal portion having an audio sampling rate, to generate anencoded audio signal, comprising: a first encoding processor configuredfor encoding the first audio signal portion in a frequency domain toobtain a first encoded signal portion; a second encoding processorconfigured for encoding the second audio signal portion in a time domainto obtain a second encoded signal portion, the second audio signalportion comprising a low band and a high band; and a controllerconfigured for analyzing a portion of the audio signal and fordetermining, that the portion of the audio signal is either the firstaudio signal portion encoded in the frequency domain or the second audiosignal portion encoded in the time domain; and an encoded signal formerconfigured for forming the encoded audio signal comprising the firstencoded signal portion for the first audio signal portion and the secondencoded signal portion for the second audio signal portion.
 2. The audioencoder of claim 1, further comprising: a preprocessor configured forpreprocessing the first audio signal portion and the second audio signalportion, wherein the preprocessor comprises: a prediction analyzerconfigured for determining prediction coefficients; and wherein thesecond encoding processor comprises: a prediction coefficient quantizerconfigured for generating a quantized version of the predictioncoefficients; and an entropy coder configured for generating an encodedversion of the quantized prediction coefficients, wherein the encodedsignal former is configured for introducing the encoded version of thequantized prediction coefficients into the encoded audio signal.
 3. Theaudio encoder of claim 1, comprising a preprocessor, wherein thepreprocessor comprises a resampler configured for resampling the audiosignal to a lower sampling rate of the second encoding processor toobtain a resampled audio signal; and wherein a prediction analyzer isconfigured to determine prediction coefficients using the resampledaudio signal.
 4. The audio encoder of claim 1, comprising apreprocessor, wherein the preprocessor comprises a long term predictionanalysis stage configured for determining one or more long termprediction parameters for the first audio signal portion.
 5. The audioencoder of claim 1, further comprising a cross-processor configured forcalculating, from an encoded spectral representation of the first audiosignal portion, initialization data of the second encoding processor, sothat the second encoding processor is initialized to encode the secondaudio signal portion immediately following the first audio signalportion in time in the audio signal.
 6. The audio encoder of claim 5,wherein the cross-processor comprises: a spectral decoder configured forcalculating a decoded version of the first encoded signal portion; aweighted prediction coefficient analysis filtering block configured forfiltering the decoded version of the first encoded signal portion toobtain a filtered decoded version and for feeding the filtered versioninto a codebook determinator of the second encoding processor for aninitialization.
 7. The audio encoder of claim 5, wherein thecross-processor comprises: a spectral decoder for calculating a decodedversion of the first encoded signal portion; and an analysis filteringstage configured for filtering the decoded version of the first encodedsignal portion or a pre-emphasized decoded version of the first encodedsignal portion to obtain a filter residual and configured for feedingthe filter residual into an adaptive codebook determinator of the secondencoding processor for an initialization.
 8. The audio encoder of claim5, wherein the cross-processor comprises: a spectral decoder configuredfor calculating a decoded version of the first encoded signal portion; apre-emphasis filter configured for filtering the decoded version of thefirst encoded signal portion to obtain a pre-emphasized version and forfeeding the pre-emphasized version or a delayed pre-emphasized versionto a synthesis filtering stage of the second encoding processor for aninitialization.
 9. The audio encoder of claim 1, wherein an analyzer isconfigured to perform a temporal tile shaping or temporal noise shapinganalysis or an operation of setting to zero spectral values in thesecond spectral portions, wherein the first encoding processor isconfigured to perform a shaping of spectral values of first spectralportions using prediction coefficients derived from the first audiosignal portion, and wherein the first encoding processor is furthermoreconfigured to perform a quantization and entropy coding operation ofshaped spectral values of the first spectral portions, and whereinspectral values of the second spectral portions are set to zero.
 10. Theaudio encoder of claim 1, wherein the second encoding processorcomprises at least one block of the following group of blocks: aprediction analysis filter; an ACELP/gain coding stage; and a predictionsynthesis filtering stage.
 11. The audio encoder of claim 1, wherein thesecond encoding processor comprises at least one block of the followinggroup of blocks: an adaptive codebook stage; an innovative codebookstage; an estimator configured for estimating an innovative codebookentry;
 12. The audio encoder of claim 1, wherein the second encodingprocessor comprises at least one block of the following group of blocks:a de-emphasis stage; and a bass post-filter analysis stage.
 13. Theaudio encoder of claim 1, wherein the time frequency converter isconfigured for converting the first audio signal portion into thefrequency domain representation comprising spectral lines up to amaximum frequency of the first audio signal portion, and wherein theanalyzer is configured for analyzing the frequency domain representationup to the maximum frequency.
 14. The audio encoder of claim 1, whereinthe second encoding processor comprises: a sampling rate converterconfigured for converting the second audio signal portion to a lowersampling rate representation of the second audio signal portion, whereinthe sampling rate converter is configured so that a lower sampling rateof the lower sampling rate representation is lower than the audiosampling rate of the second audio signal portion, and so that the lowersampling rate representation of the second audio signal portioncomprises the low band of the second audio signal portion and does notcomprise the high band of the second audio signal portion; a time domainlow band encoder configured for time domain encoding the lower samplingrate representation of the second audio signal portion; and a timedomain bandwidth extension encoder configured for parametricallyencoding the high band of the second audio signal portion.
 15. An audiodecoder for decoding an encoded audio signal comprising a first encodedaudio signal portion and a second encoded audio signal portion to obtaina decoded audio signal, comprising: a first decoding processorconfigured for decoding the first encoded audio signal portion in afrequency domain to obtain a decoded time domain first audio signalportion; a second decoding processor configured for decoding the secondencoded audio signal portion in the time domain to acquire a decodedtime domain second audio signal portion having a low band and a highband; and a combiner configured for combining the decoded time domainfirst audio signal portion and the decoded time domain second audiosignal portion to acquire the decoded audio signal.
 16. The audiodecoder of claim 15, wherein an upsampler of the second decodingprocessor comprises an analysis filterbank operating at a first samplingrate and a synthesis filterbank operating at a second sampling rate. 17.The audio decoder of claim 15, wherein the time domain low band decodercomprises a decoder and a synthesis filter configured for filtering aresidual signal using synthesis filter coefficients, wherein a timedomain bandwidth extension decoder of the second decoding processor isconfigured to upsample the residual signal to obtain an upsampledresidual signal and to process the upsampled residual signal using anon-linear operation to acquire a high band residual signal, and tospectrally shape the high band residual signal to acquire a high band ofthe decoded time domain second audio signal portion having a secondsampling rate.
 18. The audio decoder of claim 15, wherein the firstdecoding processor comprises an adaptive long term predictionpost-filter configured for post-filtering the decoded first audio signalportion, wherein the adaptive long term prediction post-filter iscontrolled by one or more long term prediction parameters comprised inthe encoded audio signal.
 19. The audio decoder of claim 15, wherein thesecond decoding processor comprises: a time domain low band decoderconfigured for decoding to obtain a low band time domain signal having afirst sampling rate; an upsampler configured for upsampling the low bandtime domain signal to obtain an upsampled low band time domain signalhaving a second sampling rate being higher than the first sampling rate,the upsampled low band time domain signal representing the low band ofthe decoded time domain second audio signal portion; a time domainbandwidth extension decoder configured for synthesizing the high band ofthe decoded time domain second audio signal portion having the secondsampling rate using the low band time domain signal; and a mixerconfigured for mixing the high band of the decoded time domain secondaudio signal portion having the second sampling rate and the upsampledlow band time domain signal having the second sampling rate to obtainthe decoded time domain second audio signal portion.
 20. The audiodecoder of claim 15, wherein the first decoding processor comprises: aspectral decoder configured for decoding first spectral portions with ahigh spectral resolution and for synthesizing second spectral portionsusing a parametric representation of the second spectral portions and atleast a decoded first spectral portion to acquire a decoded spectralrepresentation; and a frequency-time converter configured for convertingthe decoded spectral representation into a time domain to acquire adecoded time domain first audio signal portion.
 21. The audio decoder ofclaim 15, wherein the second decoding processor comprises: an adaptivecodebook synthesis stage.
 22. The audio decoder of claim 15, wherein thesecond decoding processor comprises an ACELP block configured fordecoding gains and an innovative codebook.
 23. The audio decoder ofclaim 15, wherein the second decoding processor comprises at least oneblock of the group of blocks comprising: an ACELP post-processor; and ade-emphasis stage.
 24. The audio decoder of claim 15, wherein the seconddecoding processor comprises a prediction synthesis filter.
 25. Theaudio decoder of claim 15, comprising: a cross-processor configured forcalculating, from the decoded spectral representation of the firstencoded audio signal portion, initialization data of the second decodingprocessor, so that the second decoding processor is initialized todecode the encoded second audio signal portion following in time thefirst audio signal portion in the encoded audio signal, wherein thecross-processor comprises: a frequency-time converter configured forconverting the decoded spectral representation to obtain a furtherdecoded first signal portion in the time domain; a delay stageconfigured for delaying the further decoded first signal portion toobtain a delayed version and for feeding the delayed version into ade-emphasis stage of the second decoding processor for aninitialization.
 26. The audio decoder of claim 15, comprising across-processor configured for calculating, from the decoded spectralrepresentation of the first encoded audio signal portion, initializationdata of the second decoding processor, so that the second decodingprocessor is initialized to decode the encoded second audio signalportion following in time the first audio signal portion in the encodedaudio signal, wherein the cross-processor comprises: a frequency-timeconverter configured for converting the decoded spectral representationto obtain a further decoded first signal portion in the time domain; aprediction analysis filter configured for generating a predictionresidual signal from the further decoded first signal portion or apre-emphasized further decoded first signal portion and for feeding theprediction residual signal into a codebook synthesizer of the seconddecoding processor.
 27. A method of encoding an audio signal, the audiosignal comprising a first audio signal portion and a timely subsequentsecond audio signal portion having an audio sampling rate, to generatean encoded audio signal, comprising: first encoding the first audiosignal portion in a frequency domain to obtain a first encoded signalportion; second encoding the second audio signal portion in a timedomain to obtain a second encoded signal portion, the second audiosignal portion comprising a low band and a high band; analyzing aportion of the audio signal and determining that the portion of theaudio signal is either the first audio signal portion encoded in thefrequency domain or is the second audio signal portion encoded in thetime domain; and forming the encoded audio signal comprising the firstencoded signal portion for the first audio signal portion and the secondencoded signal portion for the second audio signal portion.
 28. A methodof decoding an encoded audio signal comprising a first encoded audiosignal portion and a second encoded audio signal portion to obtain adecoded audio signal, comprising: first decoding the first encoded audiosignal portion in a frequency domain to acquire a decoded time domainfirst audio signal portion; second decoding the second encoded audiosignal portion in the time domain to acquire a decoded second timedomain audio signal portion having a low band and a high band; andcombining the decoded time domain first audio signal portion and thedecoded time domain second audio signal portion to acquire the decodedaudio signal.
 29. A non-transitory digital storage medium having storedthereon a computer program for performing, when running on a computer, amethod of encoding an audio signal, the audio signal comprising a firstaudio signal portion and a timely subsequent second audio signal portionhaving an audio sampling rate, to generate an encoded audio signal, themethod comprising: first encoding the first audio signal portion in afrequency domain to obtain a first encoded signal portion; secondencoding the second audio signal portion in a time domain to obtain asecond encoded signal portion, the second audio signal portioncomprising a low band and a high band; analyzing a portion of the audiosignal and determining that the portion of the audio signal is eitherthe first audio signal portion encoded in the frequency domain or thesecond audio signal portion encoded in the time domain; and forming theencoded audio signal comprising the first encoded signal portion for thefirst audio signal portion and the second encoded signal portion for thesecond audio signal portion.
 30. A non-transitory digital storage mediumhaving stored thereon a computer program for performing, when running ona computer, a method of decoding an encoded audio signal comprising afirst encoded audio signal portion and a second encoded audio signalportion to obtain a decoded audio signal, the method comprising: firstdecoding the first encoded audio signal portion in a frequency domain toacquire a decoded time domain first audio signal portion; seconddecoding the second encoded audio signal portion in the time domain toacquire a decoded time domain second audio signal portion having a lowband and a high band; and combining the decoded time domain first audiosignal portion and the decoded time domain second audio signal portionto acquire the decoded audio signal.